Method for transmitting demodulation reference signal for uplink data in wireless communication system, and device for same

ABSTRACT

The present disclosure provides a method for transmitting a demodulation reference signal. Specifically, the method performed by the terminal comprising: receiving, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generating a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generating a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmitting, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-Phase Shift Keying (PSK) symbol as each element of the sequence.

TECHNICAL FIELD

The present disclosure relates to a wireless communication system, and more particularly, to a method for transmitting a demodulation reference signal for an uplink data and a device for supporting the same.

BACKGROUND ART

Mobile communication systems have been generally developed to provide voice services while guaranteeing user mobility. Such mobile communication systems have gradually expanded their coverage from voice services through data services up to high-speed data services. However, as current mobile communication systems suffer resource shortages and users demand even higher-speed services, development of more advanced mobile communication systems is needed.

The requirements of the next-generation mobile communication system may include supporting huge data traffic, a remarkable increase in the transfer rate of each user, the accommodation of a significantly increased number of connection devices, very low end-to-end latency, and high energy efficiency. To this end, various techniques, such as small cell enhancement, dual connectivity, massive multiple input multiple output (MIMO), in-band full duplex, non-orthogonal multiple access (NOMA), supporting super-wide band, and device networking, have been researched.

DISCLOSURE Technical Problem

An embodiment of the present disclosure provides a method for transmitting a demodulation reference signal for an uplink data by using a low PAPR sequence.

The technical objects of the present disclosure are not limited to the aforementioned technical objects, and other technical objects, which are not mentioned above, will be apparently appreciated by a person having ordinary skill in the art from the following description.

Technical Solution

In one aspect, there is provided a method of transmitting, by a terminal, a demodulation reference signal (DMRS) for an uplink data in a wireless communication system, the method comprising receiving, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generating a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generating a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmitting, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-phase shift keying (PSK) symbol as each element of the sequence.

The length-6 sequence is determined by e^(jφ(i)π/8), and the i is an index of elements of the length-6 sequence.

A value of the φ(i) comprises (−1 −7 −3 −5 −1 3), (−7 3 −7 5 −7 −3), (5 −7 7 1 5 1), (−7 3 1 5 −1 3), (−7 −5 −1 −7 −5 5), (−7 1 −3 3 7 5) and (−7 1 −3 1 5 1).

A sequence which is cyclic shifted for the φ(i) is a same sequence with the φ(i).

A number of possible value of the φ(i) is 8⁶.

A value of an auto-correlation for the low PAPR sequence is less than a specific value.

The method further comprises applying a frequency domain spectrum shaping (FDSS) to the low PAPR sequence.

The low PAPR sequence is frequency division multiplexed (FDM) for 2 antenna ports as a Comb-2 form.

The low PAPR sequence which is used for each of the 2 antenna ports is different for each other.

In another aspect, there is provided a terminal for transmitting a demodulation reference signal (DMRS) for an uplink data in a wireless communication system, the terminal comprising a transceiver configured to transmit and receive radio signals; and a processor operatively coupled to the transceiver, wherein the processor controls to receive, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generate a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generate a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmit, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-phase shift keying (PSK) symbol as each element of the sequence.

In another aspect, there is provided an apparatus comprising one or more memories and one or more processors operatively coupled to the one or more memories, wherein the one or more processors control the apparatus to receive, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generate a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generate a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmit, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-phase shift keying (PSK) symbol as each element of the sequence.

In another aspect, there is provided one or more non-transitory computer-readable media (CRM) storing one or more instructions, wherein the one or more instructions executable by the one or more processors allow a terminal to receive, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generate a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generate a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmit, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-phase shift keying (PSK) symbol as each element of the sequence.

Advantageous Effects

According to the present disclosure, there is an effect that PAPR performance can be enhanced by using a sequence constituted by M-PSK and/or M-QAM symbols.

Effects obtainable in the present disclosure are not limited to the aforementioned effects and other unmentioned effects will be clearly understood by those skilled in the art from the following description.

DESCRIPTION OF DRAWINGS

The accompanying drawings, which are included as part of the detailed description in order to provide a thorough understanding of the present disclosure, provide embodiments of the present disclosure and together with the description, describe the technical features of the present disclosure.

FIG. 1 is a diagram illustrating one example of an NR system architecture.

FIG. 2 is a diagram illustrating one example of a frame structure in NR.

FIG. 3 illustrates one example of a resource grid in NR.

FIG. 4 is a diagram illustrating one example of a physical resource block in NR.

FIG. 5 is a diagram illustrating one example of a 3GPP signal transmitting and receiving method.

FIG. 6 is a block diagram of a wireless communication system to which methods proposed in the present disclosure may be applied.

FIG. 7 illustrates an SSB structure.

FIG. 8 illustrates SSB transmission.

FIG. 9 illustrates that a UE acquires information on DL time synchronization.

FIG. 10 illustrates a system information (SI) acquisition process.

FIG. 11 illustrates a method for informing SSB (SSB_tx) which is actually transmitted.

FIG. 12 is a diagram illustrating PAPR performance of many sequences for a case where an FDSS filter is used and a case where the FDSS filter is not used.

FIG. 13 illustrates one example of a system model and/or a procedure for a DFT-s-OFDM based system.

FIG. 14 is a diagram illustrating a flowchart of Method 1 proposed in the present disclosure.

FIG. 15 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

FIG. 16 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

FIG. 17 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

FIG. 18 illustrates one example for adaptively applying an FDSS filter proposed in the present disclosure.

FIG. 19 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

FIG. 20 is a flowchart illustrating one example of a method for generating a low PAPR sequence proposed in the present disclosure.

FIG. 21 illustrates a communication system applied to the present disclosure.

FIG. 22 illustrates a wireless device which may be applied to the present disclosure.

FIG. 23 illustrates a signal processing circuit applied to the present disclosure.

FIG. 24 illustrates another example of a wireless device applied to the present disclosure.

FIG. 25 illustrates a portable device applied to the present disclosure.

MODE FOR DISCLOSURE

Hereinafter, downlink (DL) means communication from the base station to the terminal and uplink (UL) means communication from the terminal to the base station. In downlink, a transmitter may be part of the base station, and a receiver may be part of the terminal. In downlink, the transmitter may be part of the terminal and the receiver may be part of the terminal. The base station may be expressed as a first communication device and the terminal may be expressed as a second communication device. A base station (BS) may be replaced with terms including a fixed station, a Node B, an evolved-NodeB (eNB), a Next Generation NodeB (gNB), a base transceiver system (BTS), an access point (AP), a network (5G network), an AI system, a road side unit (RSU), a robot, and the like. Further, a ‘terminal’ may be fixed or mobile and may be replaced with terms including a mobile station (UE), a mobile station (MS), a user terminal (UT), a mobile subscriber station (MSS), a subscriber station (SS) Advanced Mobile Station (WT), a Wireless Terminal (WT), a Machine-Type Communication (MTC) device, a Machine-to-Machine (M2M) device, and a Device-to-Device (D2D) device, a vehicle, a robot, an AI module, and the like.

The following technology may be used in various wireless access system including CDMA, FDMA, TDMA, OFDMA, SC-FDMA, and the like. The CDMA may be implemented as radio technology such as Universal Terrestrial Radio Access (UTRA) or CDMA2000. The TDMA may be implemented as radio technology such as a global system for mobile communications (GSM)/general packet radio service (GPRS)/enhanced data rates for GSM evolution (EDGE). The OFDMA may be implemented as radio technology such as Institute of Electrical and Electronics Engineers (IEEE) 802.11 (Wi-Fi), IEEE 802.16 (WiMAX), IEEE 802.20, Evolved UTRA (E-UTRA), or the like. The UTRA is a part of Universal Mobile Telecommunications System (UMTS). 3rd Generation Partnership Project (3GPP) Long Term Evolution (LTE) is a part of Evolved UMTS (E-UMTS) using the E-UTRA and LTE-Advanced (A)/LTE-A pro is an evolved version of the 3GPP LTE. 3GPP NR (New Radio or New Radio Access Technology) is an evolved version of the 3GPP LTE/LTE-A/LTE-A pro.

For clarity of description, the technical spirit of the present disclosure is described based on the 3GPP communication system (e.g., LTE-A or NR), but the technical spirit of the present disclosure are not limited thereto. LTE means technology after 3GPP TS 36.xxx Release 8. In detail, LTE technology after 3GPP TS 36.xxx Release 10 is referred to as the LTE-A and LTE technology after 3GPP TS 36.xxx Release 13 is referred to as the LTE-A pro. The 3GPP NR means technology after TS 38.xxx Release 15. The LTE/NR may be referred to as a 3GPP system. “xxx” means a standard document detail number. The LTE/NR may be collectively referred to as the 3GPP system. Matters disclosed in a standard document opened before the present disclosure may be referred to for a background art, terms, abbreviations, etc., used for describing the present disclosure. For example, the following documents may be referred to.

3GPP LTE

-   -   36.211: Physical channels and modulation     -   36.212: Multiplexing and channel coding     -   36.213: Physical layer procedures     -   36.300: Overall description     -   36.331: Radio Resource Control (RRC)

3GPP NR

-   -   38.211: Physical channels and modulation     -   38.212: Multiplexing and channel coding     -   38.213: Physical layer procedures for control     -   38.214: Physical layer procedures for data     -   38.300: NR and NG-RAN Overall Description     -   38.331: Radio Resource Control (RRC) protocol specification

NR Radio Access (NR)

As more and more communication devices require larger communication capacity, there is a need for improved mobile broadband communication compared to the existing radio access technology (RAT). Further, massive machine type communications (MTCs), which provide various services anytime and anywhere by connecting many devices and objects, are one of the major issues to be considered in the next generation communication. In addition, a communication system design considering a service/UE sensitive to reliability and latency is being discussed. The introduction of next generation radio access technology considering enhanced mobile broadband communication (eMBB), massive MTC (mMTC), ultra-reliable and low latency communication (URLLC) is discussed, and in the present disclosure, the technology is called new RAT for convenience. The NR is an expression representing an example of 5G radio access technology (RAT).

In a New RAT system including NR uses an OFDM transmission scheme or a similar transmission scheme thereto. The new RAT system may follow OFDM parameters different from OFDM parameters of LTE. Alternatively, the new RAT system may follow numerology of conventional LTE/LTE-A as it is or have a larger system bandwidth (e.g., 100 MHz). Alternatively, one cell may support a plurality of numerologies. In other words, UEs that operate with different numerologies may coexist in one cell.

The numerology corresponds to one subcarrier spacing in a frequency domain. Different numerology may be defined by scaling reference subcarrier spacing to an integer N.

System Architecture

FIG. 1 is a diagram illustrating one example of an NR system architecture.

Referring to FIG. 1 , NG-RAN is constituted by an NG-RA user plane (new AS sublayer/PDCP/RLC/MAC/PHY) and gNBs providing a control plane (RRC) protocol termination for a user equipment (UE). The gNBs are interconnected through an Xn interface. The gNB is also connected to NGC through an NG interface. More specifically, the gNB is connected to an Access and Mobility Management Function (AMF) through an N2 interface and a User Plane Function (UPF) through an N3 interface.

Frame Structure

FIG. 2 is a diagram illustrating one example of a frame structure in NR.

The NR system may support multiple numerologies. Here, the numerology may be defined by a subcarrier spacing and cyclic prefix (CP) overhead. In this case, multiple subcarrier spacings may be derived by scaling a basic subcarrier spacing with an integer N (or μ). Further, even if it is assumed that a very low subcarrier spacing is not used at a very high carrier frequency, the used numerology may be selected independently of a frequency band.

In addition, in the NR systems, various frame structures depending on multiple numerologies may be supported.

Hereinafter, Orthogonal Frequency Division Multiplexing (OFDM) numerology and the frame structure which may be considered in the NR system will be described.

Multiple OFDM numerologies supported in the NR systems may be defined as shown in Table 1.

TABLE 1 μ Δf = 2^(μ) · 15 [kHz] Cyclic prefix(CP) 0 15 Normal 1 30 Normal 2 60 Normal, Extended 3 120 Normal 4 240 Normal

The NR supports multiple numerologies (or subcarrier spacing (SCS)) for supporting various 5G services. For example, when the SCS is 15 kHz, a wide area in traditional cellular bands is supported and when the SCS is 30 kHz/60 kHz, dense-urban, lower latency, and wider carrier bandwidth are supported, and when the SCS is 60 kHz or higher therethan, a bandwidth larger than 24.25 GHz is supported in order to overcome phase noise.

An NR frequency band is defined as frequency ranges of two types (FR1 and FR2). FR1 may mean a sub 6-GHz range and FR2 as above 6-GHz range may mean millimeter wave (mmW).

Table 2 below shows a definition of an NR frequency band.

TABLE 2 Frequency Range Corresponding Subcarrier designation frequency range Spacing FR1  450 MHz-6000 MHz  15, 30, 60 kHz FR2 24250 MHz-52600 MHz 60, 120, 240 kHz

With respect to the frame structure in the NR system, sizes of various fields of a time domain is expressed in multiples of a time unit of T_(s)=1/(Δf_(max)·N_(f)). Here, Δf_(max)=480·10³ and N_(f)=4096. Downlink and uplink transmission is configured by a radio frame having an interval of T_(f)=(Δf_(max)N_(f)/100)·T_(s)=10 ms. Here, the radio frame is constituted by 10 subframes each having an interval of T_(sf)=(Δf_(max)N_(f)/1000)·T_(s)=1 ms. In this case, there may be one set of frames for uplink and one set of frames for downlink.

Further, transmission of uplink frame number i from the user equipment (UE) should be started earlier than the start of the downlink frame in the corresponding UE by T_(TA)=N_(TA)T_(s).

For numerology μ, slots are numbered in an increasing order of n_(s) ^(μ)ϵ{0, . . . , N_(subframe) ^(slots,μ)−1} in the subframe and numbered in an increasing order of n_(s,f) ^(μ)ϵ{0, . . . , N_(frame) ^(slots,μ)−1} in the radio frame. One slot is constituted by s_(symb) ^(μ) consecutive OFDM symbols and N_(symb) ^(μ) is determined according to the used numerology and a slot configuration. A start of slot n_(s) ^(μ) in the subframe is temporally aligned with the start of OFDM symbol n_(s) ^(μ)N_(symb) ^(μ) in the same subframe.

All UEs may not simultaneously perform transmission and reception and this means that all OFDM symbols of a downlink slot or an uplink slot may not be used.

Table 3 shows the number N_(symb) ^(slot) of OFDM symbols for each slot, the number N_(slot) ^(frame,μ) of slots for each radio frame, and the number N_(slot) ^(subframe,μ) of slots for each subframe in a normal CP and Table 4 shows the number of OFDM symbols for each slot, the number of slots for each radio frame, and the number of slots for each subframe in an extended CP.

TABLE 3 μ N_(symb) ^(slot) N_(slot) ^(frame, μ) N_(slot) ^(subframe, μ) 0 14 10 1 1 14 20 2 2 14 40 4 3 14 80 8 4 14 160 16

TABLE 4 μ N_(symb) ^(slot) N_(slot) ^(frame, μ) N_(slot) ^(subframe, μ) 2 12 40 4

FIG. 2 illustrates one example of a case where μ=2 and referring to Table 3, 1 subframe may include 4 slots. 1 subframe={1, 2, 4} slots illustrated in FIG. 2 is one example and the number of slots which may be included in 1 subframe is defined as shown in Table 3 or 4.

Further, mini-slot may include 2, 4, or 7 symbols or may include more symbols or less symbols therethan.

Physical Resource

With respect to the physical resource in the NR system, an antenna port, a resource grid, a resource element, a resource block, a carrier part, and the like may be considered.

Hereinafter, the physical resources which may be considered in the NR system will be described in detail.

First, with respect to the antenna port, the antenna port is defined so that a channel in which the symbol on the antenna port is transported may be inferred from a channel in which different symbols on the same antenna port are transported. When a large-scale property of a channel in which a symbol on one antenna port is transported may be interred from a channel in which symbols on different antenna ports are transported, two antenna ports may have a quasi co-located or quasi co-location (QC/QCL) relationship. Here, the large-scale property includes at least one of a delay spread, a Doppler spread, a frequency shift, average received power, and a received timing.

FIG. 3 illustrates one example of a resource grid in NR.

Referring to FIG. 3 , it is exemplarily described that the resource grid is constituted by N_(RB) ^(μ)N_(sc) ^(RB) subcarriers on the frequency domain and one subframe is constituted by 14·2μ OFDM symbols, but the present disclosure is not limited thereto.

In the NR system, a transmitted signal is described by one or more resource grids constituted by N_(RB) ^(μ)N_(sc) ^(RB) subcarriers and 2^(μ)N_(symb) ^((μ)) OFDM symbols. Here, N_(RB) ^(μ)≤N_(RB) ^(max,μ). The N_(RB) ^(max,μ) represents a maximum transmission bandwidth and this may vary between uplink and downlink in addition to numerologies.

In this case, as illustrated in FIG. 2 , one resource grid may be configured for each numerology μ and antenna port p.

Each resource element of the resource grid for the numerology μ and the antenna port p is referred to as the resource element and uniquely identified by an index pair (k, l). Here k=0, . . . , N_(RB) ^(μ)N_(sc) ^(RB)−1 represents an index on the frequency domain and l=0, . . . , 2^(μ)N_(symb) ^((μ))−1 refers to the position of the symbol in the subframe. When the resource element in the slot is referred to, the index pair (k, l) is used. Here l=0, . . . , N_(symb) ^(μ)−1.

A resource element for the numerology μ and the antenna port p corresponds to a complex value a_(k,l) ^((p,μ)). When there is no risk of confusion or when a specific antenna port or numerology is not specified, the indexes p and μ may be dropped, and as a result, the complex value may be a_(k,l) ^((p)) or a_(k,l) .

Further, the resource block (RB) is defined N_(sc) ^(RB)=12 consecutive subcarriers on the frequency domain.

Point A serves as a common reference point of a resource block grid and is acquired as follows.

-   -   OffsetToPointA for PCell downlink indicates the frequency offset         between the lowest subcarrier of the lowest resource block         overlapping the SS/PBCH block used by the UE for initial cell         selection and point A and is expressed by resource block units         assuming a 15 kHz subcarrier spacing for FR1 and a 60 kHz         subcarrier spacing for FR2; and     -   absoluteFrequencyPointA indicates the frequency-position of         point A expressed as in an absolute radio-frequency channel         number (ARFCN).

Common resource blocks are numbered upward from 0 in the frequency domain for a subcarrier spacing configuration μ.

A center of subcarrier 0 common resource block 0 for the subcarrier spacing configuration μ coincides with ‘point A’. A resource element (k, l) for the common resource block number n_(CRB) ^(μ) and the subcarrier spacing configuration μ in the frequency domain is given as shown in Equation 1 below.

$\begin{matrix} {n_{CRB}^{\mu} = \left\lfloor \frac{k}{N_{sc}^{RB}} \right\rfloor} & \left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack \end{matrix}$

Here, k is relatively defined at point A so that k=0 to correspond to the subcarrier centering point A. Physical resource blocks are numbered from 0 to N_(BWP,i) ^(size)−1 within a bandwidth part (BWP) and i represents the number of the BWP. A relationship between physical resource block n_(PRB) and common resource bock n_(CRB) is given by Equation D2 below.

n _(CRB) =n _(PRB) +N _(BWP,i) ^(start)  [Equation 2]

Here, N_(BWP,i) ^(start) represents a common resource block in which the BWP relatively starts to common resource block 0.

Bandwidth Part (BWP)

The NR system may support up to 400 MHz per component carrier (CC). If a UE which operates in wideband CC operates while continuously turning on RF for all CCs, UE battery consumption may increase. Alternatively, when several use cases (e.g., eMBB, URLLC, Mmtc, V2X, etc.) which operate in one wideband CC are considered, different numerologies (e.g., sub-carrier spacing) may be supported for each frequency band in the corresponding CC. Alternatively, a capability for the maximum bandwidth may vary for each UE. By considering this, the eNB may instruct the UE to operate only in a partial bandwidth rather than the entire bandwidth of the wideband CC and the corresponding partial bandwidth is defined as the bandwidth part (BWP) for convenience. The BWP may be constituted by consecutive resource blocks (RBs) on the frequency axis and may correspond to one numerology (e.g., sub-carrier spacing, CP length, slot/mini-slot duration).

Meanwhile, the eNB may configure multiple BWPs even in one CC configured to the UE. As one example, a BWP occupying a relatively small frequency domain may be configured in a PDCCH monitoring slot and PDSCH indicated in PDCCH may be scheduled onto a BWP larger therethan. Alternatively, when UEs are concentrated on a specific BWP, some UEs may be configured to other BWPs for load balancing. Alternatively, a partial spectrum of the entire bandwidth may be excluded both BWPs may be configured even in the same slot by considering frequency domain inter-cell interference cancellation between neighboring cells. In other words, the eNB may configure at least one DL/UL BWP to the UE associated with the wideband CC and activate at least one DL/UL BWP (by L1 signaling or MAC CE or RRC signaling) among configured DL/UL BWP(s) at a specific time and switching may be indicated to another configured DL/UL BWP (by L1 signaling or MAC CE or RRC signaling) or when a timer value is expired based on a timer, the timer value may be switched to the DL/UL BWP. In this case, the activated DL/UL BWP is defined as an active DL/UL BWP. However, in a situation in which the UE is in an initial access process or before RRC connection is set up, the UE may not receive a configuration for the DL/UL BWP and in such a situation, the DL/UL BWP assumed by the UE is defined as an initial active DL/UL BWP.

3GPP Signal Transmitting and Receiving Method

FIG. 5 is a diagram illustrating one example of a 3GPP signal transmitting and receiving method.

Referring to FIG. 5 , when the UE is powered on or newly enters a cell, the UE performs an initial cell search operation such as synchronizing with the eNB (S201). To this end, the UE may receive a Primary Synchronization Channel (P-SCH) and a Secondary Synchronization Channel (S-SCH) from the eNB and synchronize with the eNB and acquire information such as a cell ID or the like. Thereafter, the UE may receive a Physical Broadcast Channel (PBCH) from the eNB and acquire in-cell broadcast information. Meanwhile, the UE receives a Downlink Reference Signal (DL RS) in an initial cell search step to check a downlink channel status.

A UE that completes the initial cell search receives a Physical Downlink Control Channel (PDCCH) and a Physical Downlink Control Channel (PDSCH) according to information loaded on the PDCCH to acquire more specific system information (S202).

Meanwhile, when there is no radio resource first accessing the eNB or for signal transmission, the UE may perform a Random Access Procedure (RACH) to the eNB (S203 to S206). To this end, the UE may transmit a specific sequence to a preamble through a Physical Random Access Channel (PRACH) (S203 and S205) and receive a response message for the preamble through the PDCCH and a corresponding PDSCH (S204 and S206). In the case of a contention based RACH, a Contention Resolution Procedure may be additionally performed.

The UE that performs the above procedure may then perform PDCCH/PDSCH reception (S207) and Physical Uplink Shared Channel (PUSCH)/Physical Uplink Control Channel (PUCCH) transmission (S208) as a general uplink/downlink signal transmission procedure. In particular, the UE receives Downlink Control Information (DCI) through the PDCCH. Here, the DCI may include control information such as resource allocation information for the UE and formats may be different from each other according to a use purpose.

Meanwhile, the control information which the UE transmits to the eNB through the uplink or the UE receives from the eNB includes a downlink/uplink ACK/NACK signal, a Channel Quality Indicator (CQI), a Precoding Matrix Index (PMI), a Rank Indicator (RI), and the like. In the case of the 3GPP LTE system, the UE may transmit the control information such as the CQI/PMI/RI, etc., through the PUSCH and/or PUCCH.

Table 5 shows one example of a DCI format in the NR system.

TABLE 5 DCI format Usage 0_0 Scheduling of PUSCH in one cell 0_1 Scheduling of PUSCH in one cell 1_0 Scheduling of PDSCH in one cell 1_1 Scheduling of PDSCH in one cell

Referring to Table 5, DCI format 0_0 is used for scheduling of the PUSCH in one cell.

Information included in DCI format 0_0 is CRC-scrambled and transmitted by C-RNTI, CS-RNTI, or MCS-C-RNTI. In addition, DCI format 0_1 is used for reserving the PUSCH in one cell. Information included in DCI format 0_1 is CRC-scrambled and transmitted by C-RNTI, CS-RNTI, SP-CSI-RNTI, or MCS-C-RNTI. DCI format 1_0 is sued for scheduling of the PDSCH in one DL cell. Information included in DCI format 1_0 is CRC-scrambled and transmitted by C-RNTI, CS-RNTI, or MCS-C-RNTI. DCI format 1_1 is sued for scheduling of the PDSCH in one cell. Information included in DCI format 1_1 is CRC-scrambled and transmitted by C-RNTI, CS-RNTI, or MCS-C-RNTI. DCI format 2_1 is used to inform PRB(s) and OFDM symbol(s) of which the UE may assume not intending transmission.

The following information included in DCI format 2_1 is CRC-scrambled and transmitted by INT-RNTI.

-   -   preemption indication 1, preemption indication 2, . . . ,         preemption indication N.

Block Diagram of Wireless Communication System

FIG. 6 is a block diagram of a wireless communication system to which methods proposed in the present disclosure may be applied.

Referring to FIG. 6 , a wireless communication system includes a first communication device 910 and/or a second communication device 920. The expression of ‘A and/or B’ may be construed as the same meaning as ‘including at least one of A and B’. The first communication device may indicate the eNB and the second communication device may indicate the UE (or the first communication device may indicate the UE and the second communication device may indicate the eNB).

A base station (BS) may be replaced with terms including a fixed station, a Node B, an evolved-NodeB (eNB), a Next Generation NodeB (gNB), a base transceiver system (BTS), an access point (AP), general NB (gNB), a 5G system, a network, an AI system, a road side unit (RSU), a robot, and the like. Further, a ‘terminal’ may be fixed or mobile and may be replaced with terms including a mobile station (UE), a mobile station (MS), a user terminal (UT), a mobile subscriber station (MSS), a subscriber station (SS) Advanced Mobile Station (WT), a Wireless Terminal (WT), a Machine-Type Communication (MTC) device, a Machine-to-Machine (M2M) device, and a Device-to-Device (D2D) device, a vehicle, a robot, an AI module, and the like.

The first communication device and the second communication device include processors 911 and 921, memories 914 and 924, one or more Tx/Rx radio frequency (RF) modules 915 and 925, Tx processors 912 and 922, Rx processors 913 and 923, and antennas 916 and 926. The processor implements a function, a process, and/or a method which are described above. More specifically, a higher layer packet from a core network is provided to the processor 911 in DL (communication from the first communication device to the second communication device). The processor implements a function of an L2 layer. In the DL, the processor provides multiplexing between a logical channel and a transmission channel and allocation of radio resources to the second communication device 920 and takes charge of signaling to the second communication device. The transmit (TX) processor 912 implement various signal processing functions for an L1 layer (i.e., physical layer). The signal processing functions facilitate forward error correction (FEC) at the second communication device and include coding and interleaving. Encoded and modulated symbols are divided into parallel streams, each stream is mapped to an OFDM subcarrier, multiplexed with a reference signal (RS) in a time and/or frequency domain, and combined together by using inverse fast Fourier transform (IFFT) to create a physical channel carrying a time domain OFDMA symbol stream. An OFDM stream is spatially precoded in order to create multiple spatial streams. Respective spatial streams may be provided to different antennas 916 via individual Tx/Rx modules (or transceivers, 915). Each Tx/Rx module may modulate an RF carrier into each spatial stream for transmission. In the second communication device, each Tx/Rx module (or transceiver, 925) receives a signal through each antenna 926 of each Tx/Rx module. Each Tx/Rx module reconstructs information modulated with the RF carrier and provides the reconstructed information to the receive (RX) processor 923. The RX processor implements various signal processing functions of layer 1. The RX processor may perform spatial processing on information in order to reconstruct an arbitrary spatial stream which is directed for the second communication device. When multiple spatial streams are directed to the second communication device, the multiple spatial streams may be combined into a single OFDMA symbol stream by multiple RX processors. The RX processor transforms the OFDMA symbol stream from the time domain to the frequency domain by using fast Fourier transform (FFT). A frequency domain signal includes individual OFDMA symbol streams for respective subcarriers of the OFDM signal. Symbols on the respective subcarriers and the reference signal are reconstructed and demodulated by determining most likely signal arrangement points transmitted by the first communication device. The soft decisions may be based on channel estimation values. The soft decisions are decoded and deinterleaved to reconstruct data and control signals originally transmitted by the first communication device on the physical channel. The corresponding data and control signals are provided to the processor 921.

UL (communication from the second communication device to the first communication device) is processed by the first communication device 910 in a scheme similar to a description of a receiver function in the second communication device 920. Each Tx/Rx module 925 receives the signal through each antenna 926. Each Tx/Rx module provides the RF carrier and information to the RX processor 923. The processor 921 may be associated with the memory 924 storing a program code and data. The memory may be referred to as a computer readable medium.

Abbreviation and Definition

PUSCH: Physical Uplink Shared Channel

PUCCH: Physical Uplink Control Channel

FDSS: Frequency Domain Spectrum Shaping

PSK: Phase Shift Keying

QAM: Quadrature Amplitude Modulation

PAPR: Peak-to-Average Power Ratio

DMRS: DeModulation Reference Signals

ACK: Acknowledgement

NACK: Negative Acknowledgement

CA: Carrier aggregation

DCI: Downlink Control format Indicator/index

MAC-CE: Multiple Access Channel Control Elements

BWP: Bandwidth part

RF: Radio frequency

CC: Component carrier

SS: Synchronization Signals

SSB: Synchronization signal block—The SSB is regarded to be the same as the SS/PBCH block in the present disclosure.

SSBRI: SSB resource index/indicator

IM: Interference measurement

FDM: Frequency division multiplexing

TDM: Time division multiplexing

RS: Reference Signal(s)

CSI-RS or CSIRS: Channel State Information Reference Signals

CSI-IM: Channel State Information Interference Measurement

CRI: CSI-RS resource index/indicator

DM-RS or DMRS: Demodulation Reference Signals

MAC: Medium Access Control

MAC-CE: Medium Access Control Channel Element

NZP: Non Zero Power

ZP: Zero power

PT-RS or PTRS: Phase Tracking Reference Signals

SRS: Sounding Reference Signals

SRI: SRS resource index/indicator

PRS: Positioning Reference Signals

PRI: PRS resource index/indicator

OFDM: Orthogonal Frequency Division Multiplexing

TX: Transmitter

TP: Transmission Point

BS: Base station

RX: Receiver

RRC: Radio Resource Control

RSRP: Reference Signal Received Power

RSRQ: Reference Signal Received Quality

SNR: Signal to Noise Ratio

SINR: Signal to Interference plus Noise Ratio

URLLC: Ultra Reliable Low Latency Communication

PUSCH: Physical Uplink Shared Channels

PUCCH: Physical Uplink Control Channels

PDCCH: Physical Downlink Control Channels

PDSCH: Physical Downlink Shared Channels

ID: Identity (or meaning identity/identification number)

UL: Uplink

DL: Downlink

UE: User equipment (meaning the UE)

gNB: generic NodeB (similar concept to the eNB)

Initial Access (IA) Procedure

Synchronization Signal Block (SSB) Transmission and Related Operation

FIG. 7 illustrates an SSB structure. The UE may perform cell search, system information acquisition, beam alignment for initial access, DL measurement, etc., based on an SSB. The SSB is mixedly used with an SS/Synchronization Signal/Physical Broadcast channel (PBCH) block.

Referring to FIG. 7 , the SSB is constituted by PSS, SSS, and PBCH. The SSB is constituted by four continuous OFDM symbols and the PSS, the PBCH, the SSS/PBCH, and the PBCH are transmitted for each OFDM symbol. Each of the PSS and the SSS may be constituted by one OFDM symbol and 127 subcarriers and the PBCH is constituted by 3 OFDM symbols and 576 subcarriers. Polar coding and quadrature phase shift keying (QPSK) are applied to the PBCH. The PBCH is constituted by a data RE and a demodulation reference signal (DMRS) RE for each OFDM symbol. Three DMRS REs exist for each RB, and three data REs exist between DMRS REs.

Cell Search

The cell search refers to a process of acquiring time/frequency synchronization of the cell and detecting a cell identifier (ID) (e.g., physical layer cell ID (PCID)) of the cell by the UE. The PSS is used to detect the cell ID within a cell ID group and the SSS is used to detect the cell ID group. The PBCH is used for SSB (time) index detection and half-frame detection.

A cell search process of the UE may be organized as shown in Table 6 below.

TABLE 6 Type of Signals Operations 1st step PSS * SS/PBCH block (SSB) symbol timing acquisition * Cell ID detection within a cell ID group (3 hypothesis) 2nd Step SSS * Cell ID group detection (336 hypothesis) 3rd Step PBCH DMRS * SSB index and Half frame (HF) index (Slot and frame boundary detection) 4th Step PBCH * Time information (80 ms, System Frame Number (SFN), SSB index, HF) * Remaining Minimum System Information (RMSI) Control resource set (CORESET)/ Search space configuration 5th Step PDCCH and * Cell access information PDSCH * RACH configuration

There are 336 cell ID groups, and three cell IDs exist for each cell ID group. There may be a total of 1008 cell IDs and the cell ID may be defined by Equation 3.

N _(ID) ^(cell)=3N _(ID) ⁽¹⁾ +N _(ID) ⁽²)  [Equation 3]

Here, N_(ID) ⁽¹⁾ϵ{0, 1, . . . , 335} and N_(ID) ⁽²⁾ϵ{0, 1, 2}.

Here, NcellID represents a cell ID (e.g., PCID). N(1)ID represents a cell ID group and is provided/acquired through the SSS. N(2)ID represents a cell ID in the cell ID group and is provided/acquired through the PSS.

PSS sequence dPSS(n) may be defined to satisfy Equation 4.

d _(PSS)(n)=1 −2x(m)

m=(n+43N _(ID) ⁽²⁾)mod 127

0≤n<127

Here x(i+7)=(x(i+4)+x(i))mod 2, and

[x(6) x(5) x(4) x(3) x(2) x(1) x(0)]=[1 1 1 0 1 1 0]

SSS sequence dSSS(n) may be defined to satisfy Equation 5.

$\begin{matrix} {{d_{SSS}(n)} = {\left\lbrack {1 - {2{x_{0}\left( {\left( {n + m_{0}} \right){{mod}127}} \right)}}} \right\rbrack{\quad{{{\left\lbrack {1 - {2{x_{1}\left( {\left( {n + m_{1}} \right){mod}\mspace{11mu} 127} \right)}}} \right\rbrack m_{0}} = {{{15\left\lfloor \frac{N_{ID}^{(1)}}{112} \right\rfloor} + {5N_{ID}^{(2)}m_{1}}} = {{{N_{ID}^{(1)}\mspace{14mu}{mod}\mspace{14mu} 1120} \leq n < {127{x_{0}\left( {i + 7} \right)}}} = {\left( {{x_{0}\left( {i + 4} \right)} + {x_{0}(i)}} \right)\;{mod}\ 2{Here}}}}},{{x_{1}\left( {i + 7} \right)} = {\left( {{x_{1}\left( {i + 1} \right)} + {x_{1}(i)}} \right){mod}\ 2}},{{{and}\left\lbrack {{x_{0}(6)}\mspace{20mu}{x_{0}(5)}\mspace{20mu}{x_{0}(4)}\mspace{20mu}{x_{0}(3)}\mspace{20mu}{x_{0}(2)}\mspace{14mu}{x_{0}(1)}\mspace{14mu}{x_{0}(0)}} \right\rbrack} = {{\left\lbrack {0\mspace{14mu} 0\mspace{14mu} 0\mspace{14mu} 0\mspace{14mu} 0\mspace{14mu} 0\ 1} \right\rbrack\left\lbrack {{x_{1}(6)}\mspace{14mu}{x_{1}(5)}\mspace{20mu}{x_{1}(4)}\mspace{20mu}{x_{1}(3)}\mspace{20mu}{x_{1}(2)}\mspace{20mu}{x_{1}(1)}\mspace{20mu}{x_{1}(0)}} \right\rbrack} = {\quad{\left\lbrack {0\mspace{14mu} 0\mspace{14mu} 0\mspace{14mu} 0\mspace{14mu} 0\mspace{14mu} 0\mspace{14mu} 1} \right\rbrack.}}}}}}}} & \left\lbrack {{Equation}\mspace{14mu} 5} \right\rbrack \end{matrix}$

FIG. 8 illustrates SSB transmission.

The SSB is periodically transmitted according to SSB periodicity. An SSB basic periodicity assumed by the UE in initial cell search is defined as 20 ms. After cell access, the SSB periodicity may be configured by one of {5 ms, 10 ms, 20 ms, 40 ms, 80 ms, 160 ms} by the network (e.g., eNB). At a beginning part of the SSB periodicity, a set of SSB bursts is configured. The SSB burst set may be configured by a 5-ms time window (i.e., half-frame) and the SSB may be transmitted up to L times within the SS burst set. L which is the maximum number of transmissions of the SSB may be given as follows according to a frequency band of a carrier. One slot includes up to two SSBs.

-   -   For frequency range up to 3 GHz, L=4     -   For frequency range from 3 GHz to 6 GHz, L=8     -   For frequency range from 6 GHz to 52.6 GHz, L=64

A time position of an SSB candidate in the SS burst set may be defined as follows according to SCS. The time positions of the SSB candidates are indexed from 0 to L−1 in chronological order within the SSB burst set (i.e., half-frame).

-   -   Case A—15 kHz SCS: An index of a start symbol of the candidate         SSB is given as {2, 8}+14*n. When a carrier frequency is 3 GHz         or less, n=0, 1. When the carrier frequency is 3 to 6 GHz or         less, n=0, 1, 2, 3.     -   Case B—30 kHz SCS: The index of the start symbol of the         candidate SSB is given as {4, 8, 16, 20}+28*n. When the carrier         frequency is 3 GHz or less, n=0. When the carrier frequency is 3         to 6 GHz, n=0, 1.     -   Case C—30 kHz SCS: The index of the start symbol of the         candidate SSB is given as {2, 8}+14*n. When the carrier         frequency is 3 GHz or less, n=0, 1. When the carrier frequency         is 3 to 6 GHz or less, n=0, 1, 2, 3.     -   Case D—120 kHz SCS: The index of the start symbol of the         candidate SSB is given as {4, 8, 16, 20}+28*n. When the carrier         frequency is more than 6 GHz, n=0, 1, 2, 3, 5, 6, 7, 8, 10, 11,         12, 13, 15, 16, 17, 18.     -   Case E—240 kHz SCS: The index of the start symbol of the         candidate SSB is given as {8, 12, 16, 20, 32, 36, 40, 44}+56*n.         When the carrier frequency is more than 6 GHz, n=0, 1, 2, 3, 5,         6, 7, 8.

FIG. 9 illustrates that a UE acquires information on DL time synchronization.

The UE may acquire DL synchronization by detecting the SSB. The UE may identify the structure of the SSB burst set based on the detected SSB index, and thus detect a symbol/slot/half-frame boundary. The number of the frame/half-frame to which the detected SSB belongs may be identified using SFN information and half-frame indication information.

Specifically, the UE may acquire 10-bit System Frame Number (SFN) information from the PBCH (s0 to s9). 6 bits of the 10-bit SFN information are obtained from a Master Information Block (MIB), and the remaining 4 bits are obtained from a PBCH Transport Block (TB).

Next, the UE may acquire 1-bit half-frame indication information (c0). When a carrier frequency is 3 GHz or less, the half-frame indication information may be implicitly signaled using PBCH DMRS. The PBCH DMRS indicates 3-bit information by using one of eight PBCH DMRS sequences. Accordingly, in the case of L=4, 1 bit which remains after indicating the SSB index among 3 bits which may be indicated by using eight PBCH DRMS sequences may be used for half frame indication.

Last, the UE may acquire the SSB index based on a DMRS sequence and a PBCH payload. SSB candidates are indexed from 0 to L−1 in chronological order within the SSB burst set (i.e., half-frame). In the case of L=8 or 64, Least Significant Bit (LSB) 3 bits of the SSB index may be indicated using eight different PBCH DMRS sequences (b0 to b2). In the case of L=64, Most Significant Bit (MSB) 3 bits of the SSB index are indicated through the PBCH (b3 to b5). In the case of L=2, LSB 2 bits of the SSB index may be indicated using four different PBCH DMRS sequences (b0 and b1). In the case of L=4, 3 bits which remain after indicating the SSB index among 3 bits which may be indicated by using eight PBCH DRMS sequences may be used for the half frame indication (b2).

System Information Acquisition

FIG. 10 illustrates a system information (SI) acquisition process. The UE may acquire AS-/NAS-information through an SI acquisition process. The SI acquisition process may be applied to UEs which are in an RRC_IDLE state, an RRC_INACTIVE state, an RRC_CONNECTED state.

The SI is divided into a master information block (MIB) and a plurality of system information blocks (SIB). SI other than the MIB may be referred to as Remaining Minimum System Information (RSI). The following may be referred to for details.

-   -   The MIB includes information/parameters related to         SystemInformationBlock1 (SIB1) reception and is transmitted         through the PBCH of the SSB. In initial cell selection, the UE         assumes that the half frame with the SSB is repeated with a         periodicity of 20 ms. The UE may check whether a Control         Resource Set (CORESET) for a Type0-PDCCH common search space         exists based on the MIB. The Type0-PDCCH common search space is         a kind of PDCCH search space and is used to transmit a PDCCH for         scheduling an SI message. If there is the Type0-PDCCH common         search space, the UE may (i) a plurality of continuous RBs and         one or more continuous symbols constituting the CORESET and (ii)         a PDCCH occasion (i.e., a time domain location for receiving the         PDCCH) based on information (e.g., pdcch-ConfigSIB1) in the MIB.         If there is no Type0-PDCCH common search space, pdcch-ConfigSIB1         provides information on a frequency location where SSB/SIB1         exists and a frequency range where the SSB/SIB1 does not exist.     -   The SIB1 contains information related to the availability and         scheduling (e.g., transmission periodicity, SI-window size) of         the remaining SIBs (hereinafter, referred to as SIBx, x is an         integer of 2 or more). For example, the SIB1 may inform whether         the SIBx is periodically broadcasted or whether the SIBx is         provided by a request of the UE according to an on-demand         scheme. When the SIBx is provided by the on-demand scheme, the         SIB1 may include information which the UE requires for         performing an SI request. The SIB1 is transmitted through the         PDSCH, the PDCCH for scheduling the SIB1 is transmitted through         the Type0-PDCCH common search space, and the SIB1 is transmitted         through the PDSCH indicated by the PDCCH.     -   The SIBx is included in the SI message and transmitted through         the PDSCH. Each SI message is transmitted within a time window         (i.e., SI-window) which periodically occurs.

Channel Measurement and Rate-Matching

FIG. 11 illustrates a method for informing SSB (SSB_tx) which is actually transmitted.

In the SSB burst set, up to L SSBs may be transmitted and the numbers/positions of SSB which are actually transmitted may vary for each eNB/cell. The number/positions of SSBs which are actually transmitted are used for rate-matching and measurement and information on the actually transmitted SSB is indicated as follows.

-   -   In case related to rate-matching: The information may be         indicated through UE-specific RRC signaling or RMSI. The         UE-specific RRC signaling includes a full (e.g., length L)         bitmap in both below 6 GHz and above 6 GHz frequency ranges.         Meanwhile, the RMSI includes the full bitmap below 6 GHz and a         compression type bitmap as illustrated in FIG. 11 above 6 GHz.         Specifically, the information on the actually transmitted SSB         may be indicated by using group-bitmap (8 bits)+in-group bitmap         (8 bits). Herein, a resource (e.g., RE) indicated through the         UE-specific RRC signaling or RMSI may be reserved for SSB         transmission and the PDSCH/PUSCH may be rate-matched by         considering SSB resources.     -   In case related to measurement: When the network is in an RRC         connected mode, the network (e.g., eNB) may indicate an SSB set         to be measured in a measurement interval. The SSB set may be         indicated for each frequency layer. When there is no indication         for the SSB set, a default SSB set is used. The default SSB set         includes all SSBs in the measurement interval. The SSB set may         be indicated by using the full (e.g., length L) bitmap of the         RRC signaling. When the network is in an RRC idle, the default         SSB set is used.

The contents (NR system, frame structure, etc.) described above may be applied in combination with methods proposed in the present disclosure to be described below or may be supplemented to clarify technical features of the methods proposed in the present disclosure.

The expression of ‘A/B’ used in the present disclosure may be construed as the same meaning as A and/or B and at least one of A or B.

Several sequences having a specific length may be predefined. This may be used for transmission of uplink and/or downlink data signal/control signal/reference signal. The predefined sequence may be defined (or determined) according to several criteria including Peak-to-Average Power Ratio (PAPR) characteristics, auto-correlation characteristics, and the like.

The present disclosure proposes a method for designing a length-N(the length of the sequence is N) in which each element of the sequence is constituted by symbols such as M-Phase Shift Keying (PSK), M-Quadrature Amplitude Modulation (QAM), etc.

When a Frequency Domain Spectrum Shaping (FDSS) filter is used, it is generally known that the PAPR performance is enhanced. As an example therefor, FIG. 7 may be illustrated. For this reason, schemes are proposed, which consider the FDSS filter together in order to design a sequence constituted by pi/2 BPSK modulation symbols and a sequence constituted by M-PSK symbols.

FIG. 12 is a diagram illustrating PAPR performance of many sequences for a case where an FDSS filter is used and a case where the FDSS filter is not used.

In FIG. 12 , the FDSS filter corresponds to a time domain response of [0.28 1 0.28]. Here, [0.28 1 0.28] indicates that a side of a high center filter is cut off in the frequency domain.

In FIG. 12 , reference numeral 710 represents PAPR performance of a sequence using FDSS and reference numeral 720 represents PAPR performance of a sequence not using the FDSS.

When PAPR performance for a large number of sequences is viewed, it can be seen that the PAPR performance is enhanced in the case of using the FDSS filter.

For example, the FDSS corresponds to a time domain response of [0.28, 0.28, 1.00].

However, when the PAPR performance is viewed from the viewpoint of a specific one sequence, an optimal FDSS filter for minimizing the PAPR may vary for each sequence. However, when a different filter is used for each sequence, problems such as computation complexity and/or unnecessary implementation complexity of the eNB and the UE may occur. Further, the used filter may vary depending on UE and eNB implementation and the FDSS may not be used due to an increase in complexity or an increase in block error rate (BLER) depending on the use of the FDSS.

Accordingly, the present disclosure proposes a method for configuring (or defining) a sequence set by considering both a case of using the FDSS filter and a case of not using the FDSS filter when a length-N sequence set constituted by M-PSK or M-QAM symbols is configured (or defined or used).

Hereinafter, methods (or proposals) proposed in the present disclosure may be applied to waveform (CP-OFDM (or transform precoding disabled) and DFT-s-OFDM (transform precoding enabled)) used for DL transmission and/or UL transmission, respectively. In this regard, the UE may receive, from the eNB, information on a waveform to which the following proposed methods are to be applied through the RRC signaling.

In other words, the RRC signaling may include information on a type of waveform to be used for the DL transmission and/or UL transmission. In addition, the RRC signaling may be a configuration IE form of a reference signal (RS) to which a sequence generating method proposed below may be applied.

When the information indicating the type of waveform is included as the RS configuration IE form, the information may include fields (or parameters or information) shown in tables below.

Table 7 shows one example of a case where CP-OFDM is applied.

TABLE 7 transformPrecodingdisabled RS related parameters for CP-OFDM

Table 8 shows one example of a case where DFT-s-OFDM is applied.

TABLE 8 transformPrecodingEnabled RS related parameters for DFT-s-OFDM

The transform precoding may be used as an expression such as transformer precoder.

Further, equations and values related to pseudo-random sequence(c(i)) described below may be used for generation of the sequence and determination of an initialization value of the sequence proposed below.

Normal pseudo-random sequences are defined by a length-31 gold sequence. An output sequence c(n) having a length of M_(PN) is defined by Equation 3 below. Here, n=0, 1, . . . , M_(PN)−1.

c(n)=(x ₁(n+N _(C))+x ₂(n+N _(C)))mod 2

(n+31)=(x ₁(n+3)+x ₁(n))mod 2

x ₂(n+31)=(x ₂(n+3)+x ₂(n+2)+x ₂(n+1)+x ₂(n))mod 2  [Equation 6]

Here, N_(C)=1600 and a first m-sequence x₁(n) will be initialized to x₁(0)=1, x₁(n)=0, n=1, 2, . . . , 30.

Initialization of a second m-sequence x₂(n) is described by c_(init)=Σ_(i=0) ³⁰x₂(i)·2^(i) having a value depending on application of the sequence.

(Method 1)

K (>0) sequences (a set of K sequences) in which the length of the sequence is N (>0) and each element of the sequence is constituted by M-QAM symbols may be designed (or generated or defined) according to a rule (or condition) presented below. Here, since the sequence has a length of N, the total number of sequences which may be considered is M^(N). In other words, it may be regarded that a rule (or condition) of selecting (or choosing) a total of K(K≤M^(N)) sequences among a total of M^(N) available sequences is proposed.

{circle around (1)} Among a total of M^(N) sequences, K sequences may be chosen (or determined) to have a cross-correlation characteristic of a specific threshold or a specific level or less with each other.

{circle around (2)} Among a total of M^(N) sequences, K sequences having a low auto correlation characteristic of a specific threshold or a specific level or less with each other may be chosen (or determined). The auto-correlation value may be for a specific correlation lag and K sequences may be chosen (or determined) by considering a threshold for an auto-correlation value for one or more correlation lags.

{circle around (3)} Among a total of M^(N) sequences, K sequences having a low cyclic shift auto-correlation characteristic of a specific threshold or specific level or less may be chosen (or determined). As a more specific example, K sequences may be chosen in which a correlation between cyclic shift and not cyclic shift of +L, +L−1, +L−2, . . . , −L+1, and/or −L elements in the length-N sequence is low. The L is equal to or smaller than N−1.

In other words, when an n-th specific sequence is defined as x_(u)(n) sequences in which a value of Equation 7 below is small may be chosen.

$\begin{matrix} {{\sum\limits_{0 = 11}^{\;}{{x_{u}(n)}{x_{u}^{*}\left( {\left( {n + d} \right){mod}\mspace{11mu} N} \right)}}},{{{where}\mspace{14mu} d} = {- L}},{{- L} + 1},\ldots\;,{L - 1},L} & \left\lbrack {{Equation}\mspace{14mu} 7} \right\rbrack \end{matrix}$

{circle around (4)} Among K chosen sequences, all available cyclic shifted forms of a specific length-N sequence may be regarded as the same sequence. Accordingly, among K chosen sequences, no specific sequence is the same as an available cyclic shifted form of another sequence.

{circle around (5)} Among a total of M^(N) sequences, when a specific FDSS filter is applied to K sequences, the sequence may be chosen (or determined or defined) so as to show a low PAPR characteristic of a specific threshold or specific level (e.g., X (>0) dB) or less.

For example, the FDSS filter may be an FDSS filter corresponding to the time domain response [0.28 1 0.28].

Additionally, it may be considered that two or more multiple FDSS filters are used. Since the PAPR performance shown when applying each FDSS filter varies depending on the specific sequence, multiple FDSS filters may be used by considering that the PAPR performance shown when applying each FDSS filter varies depending on the specific sequence.

{circle around (6)} Among a total of M^(N) sequences, the sequence may be chosen (or determined or defined) so as to show a low PAPR characteristic of a specific threshold or specific level (e.g., Y(>0) dB) or less even though the FDSS filter is not used in K sequences.

{circle around (7)} Sequences of a form in which the same phase is multiplied for each sequence element in a specific length-N sequence are regarded as the same sequence without being considered as different sequences.

The reason is that when a sequence of a form in which only the phase is shifted is used, a problem occurs in distinguishing the sequence, such as whether the phase is shifted due to the channel, so it is difficult to use different sequences.

Method 1 above may be applied to a specific antenna port (e.g., specific Reference Signal (RS) antenna port) and the same rule may be applied to multiple antenna ports. Alternatively, some or all rules among the rules may be applied (or used) for each antenna by considering characteristics for each antenna port.

By considering all of the rules, a sequence may be chosen (or determined or used), which satisfies all conditions presented above and K sequences may be chosen by considering at least one of the rules.

-   -   For example, among a total of M^(N) sequences, K sequences are         chosen, which show PAPR performance of a specific         level/threshold (e.g., X dB) or less when applying the FDSS         filter and shows PAPR performance of a specific level/threshold         (e.g., Y dB) or less even when not applying the FDSS filter to         be defined (or chosen) as one sequence set.

The sequence and/or sequence set may be used for transmitting the reference and/or data by the UE/eNB.

In addition to the sequences chosen according to a condition depending on whether to apply the FDSS filter (assuming that the number of chosen sequences is K or more), K sequences having a low auto-correlation and/or cyclic-auto correlation of a specific level or less are additionally chosen to be determined as one sequence set.

K chosen sequences may be defined (or determined) as one sequence set and used by the UE and the eNB and the eNB may indicate/configure to the UE which sequence the UE is to use at a specific time. For reference, the M-PSK and M-QAM symbols mean a Phase shift keying modulation symbol in which a modulation order is M and a Quadrature Amplitude modulation symbol in which the modulator order is M.

FIG. 13 illustrates one example of a system model and/or a procedure for a DFT-s-OFDM based system.

As an application example of Method 1 above, a CP-OFDM based system and a DFT-s-OFDM based system may be considered. FIG. 13 above illustrates a procedure (or process) which may be required when Method 1 is applied to the DFT-s-OFDM based system.

In FIG. 13 above, a length-N sequence set may be configured in various forms including a sequence set configured by an integer index, a sequence set configured by binary information, and the like. Further, the FDSS may not be used due to a transmitter (UE or eNB) implementation complexity problem or a problem such as an increase in signal transmission error which is caused by using the FDSS filter. By reflecting this, the case of not applying the FDSS and a case of applying the FDSS are separately illustrated in FIG. 13 . Further, this may be selectively applied thereto. Although not illustrated in FIG. 13 above, only one of the case of using the FDSS filter and the case of not using the FDSS filter may be implemented according to implementation of the transmitter and the proposed scheme may be still usefully used even for the transmitter.

By the above proposed scheme, it is advantageous in that the sequence may be defined (or designed or chosen) and used by considering various implementation schemes of the transmitter.

(Method 1-1)

When K proposed sequences are used throughout multiple OFDM symbols in the same slot, the characteristics such as the cross-correlation/auto-correlation among K sequences need to be considered.

-   -   In two concatenated OFDM symbols, among K sequences, when two         sequences are used by using one sequence per symbol, a sequence         first used among K sequences and a sequence having a smallest         cross-correlation may be used in a next symbol. To this end, a         specific sequence and the sequence having the smallest         cross-correlation may be defined (or determined or configured)         as one pair. In other words, a specific sequence index u and a         sequence index u′ having the smallest cross-correlation may be         defined (or configured) as a pair. One example of the one pair         may be (u,u′).

Additionally, in Method 1 above, the sequence may be chosen or found while one or all of six rules {circle around (1)}, {circle around (2)}, {circle around (3)}, {circle around (4)}, {circle around (5)}, and {circle around (6)} mentioned as the rule (or condition) of selecting (or choosing) the sequence according to K(K≤M^(N)) which is a total number value of choosing the sequence. For example, it is assumed that the number of chosen sequences is 100 (i.e., if one sequence set is constituted by 100 sequences) and it is assumed that a sequence of satisfying all of the six conditions (or rules) is found. In this case, a maximum allowed cross-correlation value, a cyclic auto-correlation value, and maximum allowable PAPR values when applying the filter may be configured to specific values and when the sequence is chosen, the number of available sequences may exceed 100.

Accordingly, the number of sequences to be found may be found while fixing the number of sequences to be found and changing the conditions to stronger constraints.

With respect to Method 1 above, a flowchart for a proposed scheme (or algorithm) may be illustrated as shown in FIG. 14 . Further, respective steps of the flowchart may be simultaneously performed or independently performed. Alternatively, an order of the respective steps may be partially changed.

FIG. 14 is a diagram illustrating a flowchart of Method 1 proposed in the present disclosure.

In FIG. 14 , first, the transmitter (UE or eNB) determines N and M in order to find K (>1) sequences in which the length of each sequence is N (>1) and each element constituting the sequence is the M-PSK/M-QAM symbol (S1).

Thereafter, the transmitter 1) configures a PAPR value to be allowed when using a specific FDSS filter and a PAPR value to be allowed when not using the FDSS filter, 2) configures an allowed range/level of the cross-correlation and cyclic auto-correlation values between the sequences, and 3) configures the cyclic shifted sequence to be regarded as the same sequence, in order to determine characteristics of K sequences to be found (S2).

Thereafter, the transmitter finds a sequence that satisfies a predetermined condition by using the configured values (S3). Here, the transmitter repeats a process of choosing the sequence by changing one or more criteria among the configured conditions until K sequences are found when the number of found sequences exceeds K. Here, if K sequences are not accurately generated, K sequences may be excluded and discarded.

Thereafter, the transmitter configures or determines a length-N sequence set with the K chosen sequences (S4).

As a concrete example of the method, 30 sequences (sequence set) in which the sequence length is 6 and each element constituting the sequence is configured by 8-PSK symbols may be defined. In other words, in FIG. 13 , a case where N=6 and M=8 may be considered.

(Method 2)

Method 2 proposes that all sequences presented in Table 9 are used, in which each sequence element is constituted by 8-(Phase Shift Keying (PSK) symbols and the length is 6 or some of the sequences presented in Table 9 are to be used as uplink PUSCH and/or PUCCH DMRS sequences. The proposed sequences may be used for Discrete Fourier Transform Spread Orthogonal Frequency Division Multiplexing (DFT-S-OFDM) and/or Cyclic Prefix OFDM (CP-OFDM). In this case, since N=6 and M=8, the total number of sequences which may be considered is 8⁶. A rule of generating/choosing/using K (>0) some sequences among a total of 8⁶ sequences is as follows. In the proposal, K=30 is assumed.

A main feature of the proposed sequence is that a low PAPR characteristic of X(>0) dB or less is shown when the FDSS filter (FDSS filter in which the time-domain response is [0.28 1 0.28]) is applied and a low PAPR characteristic of Y(>0) dB is shown even when the FDSS filter is not applied. More specifically, the proposed sequence has the following characteristics. In other words, it is proposed that the UE/eNB is to use a sequence that satisfies the following characteristics/conditions.

-   -   It is characterized in that when the FDSS filter (FDSS filter in         which the time-domain response is [0.28 1 0.28]) is applied, the         PAPR is equal to or lower than approximately 2.1 [dB].     -   When the FDSS filter is not used, a sequence may be chosen and         used, in which the PAPR is equal to or lower than approximately         2.5 [dB].     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.2357) in +1 and         −1 cyclic auto-correlation.     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.4714) in +2,         +1, −1, and −2 cyclic auto-correlation.     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.80474) in +3,         +2, +1, −1, −2, and −3 cyclic auto-correlation.     -   Among K chosen sequences, all available cyclic shifted forms of         a specific length-N sequence may be regarded as the same         sequence. Accordingly, among K chosen sequences, no specific         sequence is the same as an available cyclic shifted form of         another sequence.

In other words, for example, sequence #1 of Table 9 is ‘−7 −5 −1 5 1 −5’ and ‘−5 −1 5 1 −5 −7’ which is a cyclic shift version thereof is the same sequence.

TABLE 9 PAPR [dB] PAPR [dB] index under FDSS filter without FDSS u Sequences ϕ_(u) (n) [0.28 1.0 0.28] filter 1 −7 1 −1 −7 3 7 2.1081 2.4678 2 −7 1 5 3 5 −3 2.0883 2.8683 3 −7 −5 −1 5 1 −5 1.9058 2.4184 4 −7 −5 3 −1 7 −5 2.0883 2.8683 5 −7 7 1 −3 3 7 1.9058 2.4184 6 −7 −3 −7 3 1 5 1.9850 2.2841 7 −7 −1 −5 5 3 5 1.9133 2.5031 8 −7 −3 −1 −3 7 3 1.9133 2.5031 9 −7 −3 1 −3 −5 −1 1.7809 2.8344 10 −7 −3 1 −3 7 5 1.9675 2.3452 11 −7 −3 1 −1 −7 5 1.9850 2.2841 12 −7 −3 1 5 1 −5 1.9558 2.3581 13 −7 −3 1 5 3 −3 1.9558 2.3581 14 −7 −1 3 1 3 −3 1.9941 2.8754 15 −7 −1 3 5 3 −3 1.9058 2.4184 16 −7 3 −1 3 7 −5 1.9850 2.2841 17 −7 3 −1 5 −7 −5 1.9133 2.5031 18 −7 3 1 3 7 −3 1.9091 2.5047 19 −7 3 7 5 1 5 1.7809 2.8344 20 −7 5 −7 −3 7 −5 1.7809 2.8344 21 −7 5 −5 −1 1 −1 1.9091 2.5047 22 −7 5 −3 −1 −3 1 2.0883 2.8683 23 −7 5 −1 −3 −1 3 1.9058 2.4184 24 −7 5 −1 −3 1 5 1.9558 2.3581 25 −7 5 −1 1 −1 3 1.9941 2.8754 26 −7 7 −7 −3 3 −1 1.9133 2.5031 27 −7 7 −5 −1 3 −1 1.9850 2.2841 28 −7 7 −5 1 −3 7 1.9941 2.8754 29 −7 7 −5 3 −1 7 2.0883 2.8683 30 −7 7 −5 5 −7 −3 1.7809 2.8344 31 −7 −5 −7 −3 5 1 2.1099 2.8312 32 −7 −5 −1 −7 3 1 2.1081 2.4678 33 −7 −5 3 1 −5 5 2.1140 2.4372 34 −7 −3 −5 −3 5 1 2.1099 2.8312 35 −7 −3 7 1 −1 7 2.1140 2.4372 36 −7 1 −3 5 −7 7 2.1099 2.8312 37 −7 3 −3 1 3 −5 2.1140 2.4372 38 −7 7 −1 1 5 −1 2.1140 2.4372

In ϕ_(u)(n) of Table 9, u represents the index of the sequence and n represents the element of the sequence (or index of the element). For example, when the length of the sequence is 6, n has 0, 1, 2, 3, 4, and 5 as shown in Table 10.

In Table 9, for example, when the index u is 1, ϕ₁(0), ϕ₁(1), ϕ₁(2), ϕ₁(3), ϕ₁(4), and ϕ₁(5) correspond to −7, 1, −1, −7, 3, and 7, respectively.

Table 19 above shows one example of an 8-PSK based sequence set (length-6) proposed in the present disclosure and the modulation symbols are generated by s_(u)(n)=e^(jϕ) ^(u) ^((n)π/8).

The PAPR performance is evaluated in DFT-s-OFDM having Comb-2 type DMRS for one RB (TS 38.211, TS 38.214, and TS 38.331 are referred to for Comb-2 type DMRS).

The modulation symbols are generated by

${s_{u}(n)} = {e^{\frac{j\;{\phi_{u}{(n)}}\pi}{8}}.}$

-   -   u: Sequence index)     -   n: Element index of each sequence)

The applied FDSS filter corresponds to the time domain response of [0.28 1.0 0.28].

-   -   An IFFT size is 64 and a DFT size is 12.

Excellence in terms of the PAPR performance of the length-6 8-PSK sequence presented in Table 9 above may be confirmed in FIG. 150 . The PAPR performances of a total of 30 sequences corresponding to #1 to #30 among the sequences presented in Table 9 are shown. FIG. 15 above shows the PAPR performance when the sequence presented in Table 9 above is used as the Comb-2 DMRS sequence in the DFT-S-OFDM system.

The PAPR performance when applying the FDD filter (corresponding to the time-domain response of [0.28 1.0 0.28]) and the PAPR performance when not applying the FDSS filter for the presented sequence may be confirmed.

The PAPR performed is presented by distinguishing whether to apply the FDSS similarly even to the conventional presented length-6 8-PSK. Both the conventional scheme and the proposed scheme consider that one sequence set is constituted by 30 sequences and are a PAPR evaluation result therefor. Accordingly, a graph may not be presented with probability (PAPR>PAPR_0)=0.1 or less due to the lack of evaluation samples, but a difference in performance between individual sequences may be clearly confirmed. As the conventional presented length-6 8-PSK sequence, sequences presented in R1-1813445, R1-190081, R1-1900020, and R1-1900673 are referred to.

As illustrated in FIG. 15 , the proposed sequence shows a performance similar to the PAPR characteristic shown by the conventional presented sequence when applying the FDSS filter. In particular, it can be seen that a fine but slightly better performance is shown in a region in which probability (PAPR>PAPR_0)=0.1 or less. When the FDSS filter is not applied, the PAPR performance deteriorates compared with the FDSS filter is applied, but a more excellent PAPR performance than the conventional presented sequence may be confirmed. In other words, it may be confirmed that in 30 proposed sequences, the PAPR does not exceed 2.5 dB even when the FDSS filter is not applied unlike the conventional presented sequence set (30 sequences are presented as one sequence set in respective references R1-1813445, R1-190081, R1-1900020, and R1-1900673).

FIG. 15 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

The PAPR values presented in Table 9 and FIG. 15 above may be slightly different depending on the IFFT size and a tool for performing a simulation, but a large trend will be similar. Therefore, it may be considered that even if there is the slight difference, the slight difference does not deviate from the spirit of the method presented in the present disclosure and included in the method proposed in the present disclosure. Further, it should be considered that a sequence generating/choosing method which exceeds the auto-correlation threshold is also included in the spirit of the method proposed in the present disclosure if the sequence is chosen/used by considering whether to use the FDSS or not.

Method 2 above may primarily adopt FDSS filter (corresponding to time-domain response [0.28, 1.00, 0.28]), but may be considered as a sequence providing a comparable PAPR performance even when the FDSS filter is not used.

On the contrary, the sequence may be used so that the PAPR performance when the FDSS filter is not used is more excellent than the PAPR performance when a specific FDSS filter is used.

However, it is designed that even when the specific FDSS filter is used, the comparable PAPR performance may be shown. If there are many cases in which the FDSS filter is not used due to an increase in error rate, which is caused by distortion of an original transmission signal due to transmitter implementation complexity and/or the use of the FDSS filter, a sequence of a scheme proposed in Method 3 below may be usefully used.

(Method 2-2)

Method 2-2 relates to a method in which all sequences presented in Table 7 are used, in which each sequence element is constituted by 8-(Phase Shift Keying (PSK) symbols and the length is 6 or some of the sequences presented in Table 7 are used as uplink PUSCH and/or PUCCH DMRS sequences. The proposed sequences may be used for Discrete Fourier Transform Spread Orthogonal Frequency Division Multiplexing (DFT-S-OFDM) and/or Cyclic Prefix OFDM (CP-OFDM).

In the case of the method, since N=6 and M=8, the total number of sequences which may be considered is 8⁶. In this example, a total of two antenna ports (e.g., two DMRS antenna ports) are considered and each antenna port is subject to Frequency Division Multiplexing (FDM) in the form of Comb-2. A rule of generating/choosing/using K (>0) some sequences among a total of 8⁶ sequences is as follows. In the method, K=45 is assumed.

A main feature of the proposed sequence is that two antenna ports subject to FDM in the form of Comb-2 show a low PAPR characteristic of 2.1 dB or less when the FDSS filter (FDSS filter in which the time-domain response is [0.28 1 0.28]) is applied and a low PAPR characteristic of 2.3 dB even when the FDSS filter is not applied.

Additionally, the proposed sequence has the following characteristics.

-   -   The proposed sequence has a characteristic that the maximum         cyclic auto-correlation is equal to or smaller than         approximately 0.2357 in +1 and −1 cyclic auto-correlation lags.     -   The proposed sequence has a characteristic that the maximum         cyclic auto-correlation is equal to or smaller than         approximately 0.8 in +3, +2, +1, −1, −2, and −3 cyclic         auto-correlation lags.     -   Among K chosen sequences, all available cyclic shifted forms of         a specific length-N sequence may be regarded as the same         sequence. Accordingly, among K chosen sequences, no specific         sequence is the same as an available cyclic shifted form of         another sequence.

In other words, for example, sequence #1 of Table 10 is ‘−7 −5 −1 5 1 −5’ and ‘−5 −1 5 1 −5 −7’ which is a cyclic shift version thereof is the same sequence.

TABLE 10 PAPR PAPR PAPR [dB] [dB] [dB] under under without FDSS FDSS FDSS filter filter filter index for the for the for both u Sequences ϕ_(u) (n) 1^(st) port 2^(nd) port ports 1. −7 −7 −5 1 −5 3 1.8602 2.0111 2.2792 2. −7 −5 −5 1 −7 3 1.8544 2.0157 2.2830 3. −7 −5 −1 −7 −5 5 1.5917 1.1343 1.8125 4. −7 −3 −7 −3 5 1 1.1298 1.5228 1.6323 5. −7 −3 −5 −1 5 1 1.5775 1.8365 2.1090 6. −7 −3 −5 −1 7 1 1.8403 1.2296 1.4634 7. −7 −3 −5 −1 7 3 1.5775 1.8365 2.1090 8. −7 −3 1 −5 5 3 1.6229 1.8270 2.1422 9. −7 −3 1 −5 7 3 1.3980 1.8686 1.4299 10. −7 −3 1 −3 7 3 1.2715 1.8500 2.2025 11. −7 −3 1 7 3 −1 1.4188 1.8577 1.3790 12. −7 −1 −5 3 7 5 1.5775 1.8365 2.1090 13. −7 −1 1 −7 3 1 1.9140 2.0557 1.9783 14. −7 1 −5 1 3 3 1.8544 2.0157 2.2830 15. −7 1 −3 1 5 1 1.8693 1.7879 2.0393 16. −7 1 −3 3 7 5 1.5775 1.8365 2.1090 17. −7 1 3 3 −3 3 2.0740 1.9021 2.0493 18. −7 1 5 1 −3 1 1.8762 1.7764 1.9936 19. −7 3 −5 −1 −3 1 1.8318 1.2377 1.4984 20. −7 3 −5 1 1 3 1.8602 2.0111 2.2792 21. −7 3 −3 −5 −1 3 1.9971 1.5769 1.7233 22. −7 3 1 −7 −3 1 1.7805 1.5540 1.8564 23. −7 3 1 −5 −1 3 1.9405 1.4918 2.0507 24. −7 3 1 −5 1 3 1.8712 1.8673 1.1808 25. −7 3 1 5 −1 3 1.7987 1.8941 2.2224 26. −7 3 3 5 −3 3 2.0908 1.9001 2.0476 27. −7 3 5 −1 3 5 1.5780 1.1347 1.8558 28. −7 3 5 7 1 −3 1.6121 1.9126 1.2472 29. −7 5 −7 −3 3 −1 1.2715 1.8500 2.2025 30. −7 5 −5 −3 3 −1 1.4544 2.0850 1.7632 31. −7 5 −5 −1 −3 1 1.5769 1.8271 2.1872 32. −7 5 −5 −1 3 −1 1.3007 1.8495 2.2771 33. −7 5 −3 1 −3 1 1.1521 1.5386 1.6826 34. −7 5 −3 1 −1 3 1.5886 1.8200 2.1604 35. −7 5 −1 −5 −1 3 1.3064 1.8552 2.2758 36. −7 5 −1 1 3 −3 1.6196 1.9058 1.2817 37. −7 5 1 −7 −3 1 1.6825 1.7349 1.5684 38. −7 5 1 −5 −1 3 1.4188 1.8577 1.3790 39. −7 5 1 5 −7 −1 2.0376 1.7490 1.8846 40. −7 5 1 7 −5 −1 1.3980 1.8686 1.4299 41. −7 7 −5 1 −3 5 1.5886 1.8200 2.1604 42. −7 7 −5 3 −1 5 1.5769 1.8271 2.1872 43. −7 7 1 −7 −5 1 1.9270 2.0557 1.9546 44. −7 7 1 −7 −3 1 1.7795 1.5824 1.8210 45. −7 7 1 −5 −1 3 1.6275 1.8289 2.0924

Table 10 shows one example of an 8-PSK based sequence set (length-6) proposed in the present disclosure. Here, the modulation symbols are generated by s_(u)(n)=e^(jϕ) ^(u) ^((n)π/8). The PAPR performance is evaluated in DFT-s-OFDM having Comb-2 type DMRS for one RB (TS 38.211, TS 38.214, and TS 38.331 are referred to for Comb-2 type DMRS).

-   -   The modulation symbols are generated by

${s_{u}(n)} = {e^{\frac{j\;{\phi_{u}{(n)}}\pi}{8}}.}$

-   -   u: Sequence index)     -   n: Element index of each sequence).     -   The applied FDSS filter corresponds to a time domain response of         [0.28 1.0 0.28].     -   An IFFT size is 64 and a DFT size is 12.

It is proposed that some or all of the sequences presented in Table 10 above are used. Further, some or all of the sequences presented in Table 10 above and sequences (having different characteristics) not presented in the above table and one sequence set may be constituted and used. It may be considered that such a constitution as an extension (or application) of the present disclosure is included in the spirit of the method proposed in the present disclosure.

In FIG. 16 , the excellence of the sequence proposed in Method 2-2 above may be confirmed. Compared with the conventional sequence, the proposed sequence shows the low PAPR characteristic in both the case of using the FDSS filter and the case of not using the FDSS filter. In particular, in the case of not using the FDSS filter, a difference in PAPR performance between the sequence presented in Method 2-2 and the conventional sequence is significantly large.

FIG. 16 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

In Methods 2 and 2-1 above, a length-6 DMRS sequence (Pre-DFT length-6 sequence) is mapped to a time-frequency resource element (RE) by the following scheme.

It is characterized in that when frequency-RE is used as Comb-2 type in the frequency domain, a time-axis signal is shown repeatedly twice. Accordingly, in order to transmit the length-6 DMRS sequence in a step before DFT in the DFT-spread-OFDM system, the length-6 sequence needs to be used by considering that the length-6 sequence is repeated twice in the time domain. Accordingly, in a (pre-)DFT procedure, the length-6 sequence should be used to be repeated twice.

$\begin{matrix} {x_{2N} = {D_{DFT}\begin{bmatrix} s_{N} \\ s_{N} \end{bmatrix}}} & \left\lbrack {{Equation}\mspace{14mu} 8} \right\rbrack \end{matrix}$

Here, D_(DFT) represents a Discrete Fourier Transform (DFT) matrix.

s_(N)=[s₁, s₂, . . . , s₆] represents a 6×1 vector (one length-6 sequence). Here, each element s_(i) represents M-PSK/M-QAM symbol.

x_(2N) represents a 12×1 vector which is a frequency-domain signal after DFT processing of

$\begin{bmatrix} s_{N} \\ s_{N} \end{bmatrix}.$

In this case, when the DMRS sequence is transmitted by the DFT-spread-OFDM scheme, a DFT computation is performed by using a form in which the length-6 sequence is repeated twice in the pre-DFT step as mentioned above at the time of transmitting a single-port DMRS.

When the number of DMRS ports is 2, a specific DMRS port may be configured in a comb-2 type in which a frequency offset is 0 and the other DMRS port may be configured in a comb-2 type in which the frequency offset is 1. In this case, when the length-6 sequence to be used in the second DMRS port and the length-6 sequence to be used in the first DMRS port are mapped to a frequency axis through a (pre-)DFT process as shown in Equation 8 above, both the length-6 sequences are allocated to a Comb-2 structure (comb-2 structure in which the frequency offset is 0) having frequency offset “0”.

When the length-6 sequence to be used in the second DMRS port is mapped to the frequency axis through the (pre-)DFT process as shown in Equation 8 above, the length-6 sequence is allocated to the Comb-2 structure having frequency offset “0”. The length-6 sequence is allocated according to Equation 9 below instead of Equation 8 in order to allocate the length-6 sequence to the Comb-2 structure having frequency offset “1” so as to prevent the first DMRS port and the RE from being overlapped with each other.

$\begin{matrix} {{x_{2N} = {K{D_{DFT}\begin{bmatrix} s_{N} \\ s_{N} \end{bmatrix}}}}{K = \begin{bmatrix} \; & \; & \; & \; & \; & 0_{1 \times 12} & \; & \; & \; & \; & \; & \; \\ 1 & 0 & 0 & 0 & 0 & \begin{matrix} 0 & 0 \end{matrix} & \; & 0 & 0 & 0 & 0 & 0 \\ \; & \; & \; & \; & \; & 0_{1 \times 12} & \; & \; & \; & \; & \; & \; \\ 0 & 0 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\ \; & \; & \; & \; & \; & 0_{1 \times 12} & \; & \; & \mspace{11mu} & \; & \; & \; \\ 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\ \; & \; & \; & \; & \; & 0_{1 \times 12} & \; & \; & \; & \; & \; & \; \\ 0 & 0 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 0 \\ \; & \; & \; & \; & \; & 0_{1 \times 12} & \; & \; & \; & \; & \; & \; \\ 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 \\ \; & \; & \; & \; & \; & 0_{1 \times 12} & \; & \; & \; & \; & \; & \; \\ 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 1 & 0 \end{bmatrix}}} & \left\lbrack {{Equation}\mspace{14mu} 9} \right\rbrack \end{matrix}$

In Equation 9 above, K is a 12×12 matrix and a matrix which allows elements allocated to an odd-numbered subcarrier resource element (RE) to be allocated to an even-numbered subcarrier RE.

(Method 3)

Method 3 relates to a method in which all sequences presented in Table 11 are used, in which each sequence element is constituted by 12-(Phase Shift Keying (PSK) symbols and the length is 6 or some of the sequences presented in Table 11 are used as uplink PUSCH and/or PUCCH DMRS sequences. The proposed sequences may be used for Discrete Fourier Transform Spread Orthogonal Frequency Division Multiplexing (DFT-s-OFDM) and/or Cyclic Prefix OFDM (CP-OFDM).

In the case of the method, since N=6 and M=8, the total number of sequences which may be considered is 8⁶.

A rule of generating (or choosing or using) K (>0) some sequences among a total of 8⁶ sequences is as follows.

A main feature of the sequence proposed in the present disclosure is that the low PAPR characteristic of Y (>0) or less is shown when the FDSS filter is not used and the PAPR performance is less excellent when the FDSS filter (FDSS filter in which the time-domain response is [0.28 1 0.28]) than the PAPR performance when the FDSS filter is not used, but the low PAPR characteristic of X (>0) dB or less is still shown. More specifically, the proposed sequence has the following characteristics. In other words, it is proposed that the UE/eNB is to use a sequence that satisfies the following characteristics/conditions.

-   -   It is characterized in that when the FDSS filter (FDSS filter in         which the time-domain response is [0.28 1 0.28]) is applied, the         PAPR is equal to or lower than approximately 2.8 [dB].     -   When the FDSS filter is not used, a sequence may be chosen and         used, in which the PAPR is equal to or lower than approximately         2.1 [dB].     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.2357) in +1 and         −1 cyclic auto-correlation.     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.4714) in +2,         +1, −1, and −2 cyclic auto-correlation.     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.80474) in +3,         +2, +1, −1, −2, and −3 cyclic auto-correlation.     -   Among K chosen sequences, all available cyclic shifted forms of         a specific length-N sequence may be regarded as the same         sequence. Accordingly, among K chosen sequences, no specific         sequence is the same as an available cyclic shifted form of         another sequence.

TABLE 11 PAPR [dB] PAPR [dB] Sequence under FDSS filter without FDSS index u Sequences ϕ_(u) (n) [0.28 1.0 0.28] filter 1. −7 −7 −5 1 −5 5 2.2775 1.7169 2. −7 −7 3 −7 −5 −1 2.5231 2.0790 3. −7 −5 −7 3 −3 5 2.5217 1.9843 4. −7 −5 −5 −3 5 1 2.5607 2.0547 5. −7 −5 −5 −1 −7 3 2.2728 1.7325 6. −7 −5 −1 7 1 −5 2.5422 2.0851 7. −7 −5 1 −5 −7 5 2.7798 1.9549 8. −7 −5 1 −5 −5 5 2.5223 2.0798 9. −7 −5 3 −1 7 −7 2.5529 2.0873 10. −7 −3 −7 7 1 7 2.7798 1.9549 11. −7 −3 −3 −1 5 −1 2.2728 1.7325 12. −7 −3 −1 −3 7 1 2.5370 2.0877 13. −7 −3 −1 5 −1 −3 2.7821 1.8653 14. −7 −3 −1 5 −1 −1 2.5344 1.9917 15. −7 −3 5 −1 −7 7 2.5217 1.9843 16. −7 −3 7 1 7 −7 2.2775 1.7169 17. −7 −3 7 7 1 7 2.5223 2.0798 18. −7 −1 −7 −7 3 7 2.5231 2.0790 19. −7 −1 −7 3 7 7 2.2728 1.7325 20. −7 −1 1 1 5 −1 2.2775 1.7169 21. −7 −1 1 5 −1 −1 2.5344 1.9917 22. −7 −1 1 5 1 −1 2.7821 1.8653 23. −7 1 −5 5 3 5 2.5370 2.0877 24. −7 1 −3 5 7 7 2.5607 2.0547 25. −7 1 3 3 5 −3 2.5529 2.0873 26. −7 1 5 7 5 −1 2.5422 2.0851 27. −7 3 −7 −5 −1 −5 2.7798 1.9549 28. −7 3 −5 −1 1 −1 2.5217 1.9843 29. −7 3 −3 −5 −3 1 2.5422 2.0851 30. −7 3 −3 1 1 3 2.2775 1.7169 31. −7 3 −3 3 5 5 2.2728 1.7325 32. −7 3 1 −3 1 3 2.7821 1.8653 33. −7 3 1 3 7 −1 2.5217 1.9843 34. −7 3 3 −3 1 3 2.5344 1.9917 35. −7 3 3 −3 3 5 2.5344 1.9917 36. −7 5 −3 −1 −1 1 2.5529 2.0873 37. −7 5 3 −3 3 5 2.7821 1.8653 38. −7 7 1 −5 3 7 2.5422 2.0851 39. −7 7 3 7 −7 −1 2.7798 1.9549

Table 11 shows one example of a proposed 8-PSK based sequence set (length-6). The modulation symbols are generated by s_(u)(n)=e^(jϕ) ^(u) ^((n)π/8). The PAPR performance is evaluated in the DFT-s-OFDM system having the Comb-2 type DMRS for one RB.

FIG. 17 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

An advantage of the length-6 8-PSK sequence presented in Table 11 above may be seen in FIG. 17 . FIG. 17 above illustrates the PAPR performance when the sequence presented in Table 11 above is used as the Comb-2 DMRS sequence in the DFT-S-OFDM system. The PAPR performance when applying the FDD filter (corresponding to the time-domain response of [0.28 1.0 0.28]) and the PAPR performance when not applying the FDSS filter for the presented sequence are shown. It may be confirmed that the sequence defined according to Method 3 above shows the PAPR performance of 2.1 dB or less when the FDSS filter is not used.

Methods 2 and 3 above relate to a method that uses a sequence which shows the low PAPR characteristic in a case of applying the FDSS filter (in particular, an FDSS filter in which the time-domain response corresponds to [0.28 1.00 0.28]) and in a case of not applying the FDSS filter, i.e., both cases with respect to the same specific sequence.

However, when the sequence having the low PAPR characteristic is determined (or used) by considering only the case where the FDSS filter is not used or the sequence having the low PAPR characteristic is determined (or used) by assuming only the FDSS filter, a sequence having a lower PAPR characteristic than the sequence in the above case may be used. In other words, since there is a limit that the PAPR performance is good in both the case of applying the filter and the case of not applying the filter, the following method is proposed by considering that there is the limit that the PAPR performance is good.

(Method 4)

Method 4 relates to a method in which elements constituting one sequence are M-PSK/M-QAM symbols and one sequence set is constituted by a total of K sequences having a sequence length of N and K₁(>0) sequences in which a specific PAPR has an excellent characteristic are used when a specific FDSS filter is used and K₂(>0) sequences having the excellent PAPR characteristic are used when the FDSS filter is not used.

The total number of sequences constituting one sequence set is K=K₁+K₂.

By considering multiple environments such as whether specific FDSS filters are used a lot or whether the FDSS filter is not primarily used, the K₁ and K₂ may be considered. For example, if K length-N sequences are defined (or constituted or used) by considering that the specific FDSS filter is primarily used, a sequence set of K₁>>K₂>0 may be constituted.

Alternatively, if one sequence set is constituted (or determined) (K sequences are constituted (or determined) by targeting the case where the FDSS filter is not used, a sequence set of K₂>>K₁>0 may be constituted (or determined). For example, by assuming K=30, an extreme case such as a case of K₁=27, K₂=3 may be considered.

-   -   For example, if 30 sequences constitute one sequence set by         considering the length-6 sequence constituted by 8-PSK symbols,         a sequence presented in Table 12 below may be used. It is         proposed that some or all of the sequences are used as a         reference signal sequence such as the DMRS in the DFT-s-OFDM         system.     -   Table 12 below is configured by assuming K=K₁+K₂=47.

As shown in Table 12, sequences #1 to #15 are sequence to use not using the FDSS filter as a main target and sequences #16 to #30 are sequences which will consider that FDSS filters corresponding to the time-domain response [0.28 1.0 0.28] are together used and uses together using the FDSS filters as a target. A PAPR performance difference depending on whether to use the filter may be confirmed in Table 13 presented. Further, the following sequence is configured to have the following characteristics.

-   -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.2357) in +1 and         −1 cyclic auto-correlation.     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.4714) in +2,         +1, −1, and −2 cyclic auto-correlation.     -   It is characterized in that the maximum cyclic auto-correlation         is low (equal to or smaller than approximately 0.80474) in +3,         +2, +1, −1, −2, and −3 cyclic auto-correlation.     -   Among K chosen sequences, all available cyclic shifted forms of         a specific length-N sequence may be regarded as the same         sequence. Accordingly, among K chosen sequences, no specific         sequence is the same as an available cyclic shifted form of         another sequence.     -   It is configured so that there is no sequence of a form in which         only the phase is just shifted among K chosen sequences. In         other words, sequences of a form in which six sequence elements         are multiplied by the same phase are regarded as the same         sequence and excluded. In Table 12 below, a first sequence         element of each sequence is fixed to −7, but when all of 6         elements corresponding to each sequence should are similarly         phase-shifted, each sequence should be regarded as the same         sequence. In other words, although not presented in Table 12         below, it can be seen that when the following sequence is         phase-shifted, each sequence which is the same sequence is         included in the sequence proposed in the present disclosure.

TABLE 12 PAPR [dB] PAPR [dB] Sequence under FDSS filter without FDSS index u Sequences ϕ_(u) (n) [0.28 1.0 0.28] filter 1. −7 −7 −5 1 −5 5 2.2775 1.7169 2. −7 −7 −5 1 −5 7 3.2781 1.8879 3. −7 −5 −5 −1 −7 3 2.2728 1.7325 4. −7 −5 7 1 7 −7 3.2781 1.8879 5. −7 −3 −3 −1 5 −1 2.2728 1.7325 6. −7 −3 −1 5 −1 −3 2.7821 1.8653 7. −7 −3 7 1 7 −7 2.2775 1.7169 8. −7 −1 −7 3 7 7 2.2728 1.7325 9. −7 −1 1 1 3 −1 3.2781 1.8879 10. −7 −1 1 1 5 −1 2.2775 1.7169 11. −7 −1 1 5 1 −1 2.7821 1.8653 12. −7 3 −3 1 1 3 2.2775 1.7169 13. −7 3 −3 3 5 5 2.2728 1.7325 14. −7 3 −1 1 1 3 3.2781 1.8879 15. −7 3 1 −3 1 3 2.7821 1.8653 16. −7 −3 −7 −3 −5 5 1.5945 3.4043 17. −7 −3 −7 −3 1 −5 1.5945 3.4043 18. −7 −3 −7 −3 1 −1 1.4462 3.1777 19. −7 −3 −7 5 −7 −3 1.5120 3.4679 20. −7 −3 −5 −1 3 −1 1.4415 3.6409 21. −7 −3 1 −3 −7 −3 1.5311 3.5024 22. −7 −3 1 −3 1 −5 1.4597 3.1244 23. −7 −3 1 −3 1 −3 1.5120 3.4679 24. −7 −3 1 −3 7 −5 1.4371 3.5745 25. −7 −3 1 −1 3 −3 1.4300 3.5842 26. −7 −3 1 5 1 −3 0.7839 3.0103 27. −7 −3 7 3 7 −5 1.4300 3.5842 28. −7 −3 7 5 −7 −3 1.4597 3.1244 29. −7 −3 7 5 −7 5 1.5945 3.4043 30. −7 3 −1 3 7 5 1.4415 3.6409 31. −7 3 1 5 −7 5 1.4462 3.1777 32. −7 3 7 3 7 −5 1.4609 3.1778 33. −7 3 7 5 −7 −3 1.4371 3.5745 34. −7 5 −7 −3 −7 5 1.5120 3.4679 35. −7 5 −7 −3 −5 −1 1.4371 3.5745 36. −7 5 −7 −3 −5 5 1.4597 3.1244 37. −7 5 −7 −3 1 −3 0.7788 2.9581 38. −7 5 −7 5 1 5 1.5311 3.5024 39. −7 5 −1 3 1 5 1.4300 3.5842 40. −7 5 1 −3 1 5 0.7839 3.0103 41. −7 5 1 5 −7 −3 0.7788 2.9581 42. −7 5 1 5 1 5 1.5120 3.4679 43. −7 7 −5 −1 −5 −1 1.4609 3.1778 44. −7 7 −5 −1 −5 5 1.4300 3.5842 45. −7 7 −5 5 1 5 1.4371 3.5745 46. −7 7 1 5 −7 5 1.5945 3.4043 47. −7 7 1 5 1 5 1.4597 3.1244

Table 12 shows one example of the proposed 8-PSK based sequence set (length-6) and the modulation symbols are generated by s_(u)(n)=e^(jϕ) ^(u) ^((n)π/8). The PAPR performance is evaluated in the DFT-s-OFDM system having the Comb-2 type DMRS for one RB.

In Table 12 above, sequences #1 to #15 are sequence to use not using the FDSS filter as a main target and sequences #16 to #47 are sequences which choose (or determine or configure) the case of using the FDSS filter corresponding to the time-domain response [0.28 1.0 0.28] as the target.

(Method 5)

In Method 5, it is apparent that when the FDSS filter is used, the PAPR characteristic is improved with probability, but the PAPR is not optimized in terms of each sequence. As described in the present disclosure, it can be seen that a specific FDSS filter is applied to a specific sequence, the PAPR characteristic of the specific sequence deteriorates.

Based thereon, it is proposed that the transmitter uses different FDSS filters according to a sequence, a sequence group, and/or a set of sequences (sub set) transmitted. Further, one sequence set optimized to the used FDSS filter is constituted (or determined) to be used by the UE/eNB. As one example, a system/device illustrated in FIG. 19 may be considered. Different sequences may be used for each specific RB(s) and/or for each a group of RB(s) and the transmitter may use different sequences for each OFDM symbol and/or for each slot(s).

In order for a receiver to help approximately reconstructing different sequences, the transmitter may inform the receiver of which FDSS filter is to used in the specific sequence. In addition/alternatively, the receiver may indicate to the transmitter which filter is to be used for each specific sequence and for each sequence group. Alternatively, the receiver and the transmitter may predefine (or promise) which FDSS filter is used for which sequence and use the corresponding FDSS filter.

-   -   One sequence set/group/table is constituted and sequences may be         used, which are optimized to FDSS filter #1, FDSS filter #2, . .         . , FDSS filter #D used. In other words, D optimized sequence         groups/sub-sets are configured, to which the PAPR performance         (and/or including various criteria/performances such as         cross-correlation/auto-correlation) is optimized by using the         FDSS filter with respect to D FDSS filters to be used and         defined (or configured) as one set/group, and as a result, the         transmitter may be used for transmitting the reference signals         including the DMRS, the SRS, the CSI-RS, and the like.     -   Additionally, by considering even the case of not using the FDSS         filter, a total of (D+1) sequence groups (or sub-sets) in which         the PAPR performance is optimized are constituted with respect         to the case of using D FDSS filters and the case of not using         the FDSS filter and one sequence set/group/table is constituted,         and as a result, the transmitter may be used for transmitting         the reference signal/data signal.

FIG. 18 illustrates one example for adaptively applying an FDSS filter proposed in the present disclosure.

NR Rel-16 supports binary CGS in Tables 1, 2, and 3 with respect to lengths 12, 18, and 14, respectively and then pi/2 BPSK modulation is followed, and then DFT is followed as a DMRS sequence for π/2 BPSK modulation with respect to both the PUSCH and the PUCCH.

The above contents are applicable to a single DMRS configuration. CGS for a 2-symbol DMRS configuration may be discussed. Tables 1, 2, and 3 may be discovered in R1-1901362.

Here, the CGS may be used up to length-8 and 8-PSK may be used for length-6.

Hereinafter, in the 2-symbol DMRS configuration, a computer generation sequence applying method will be described.

In two symbols, the same computer generate sequence (CGS) may be used.

In a CGS set (among 30 sequences), when a specific sequence is used in a first symbol, the specific sequence is configured to be used in a next symbol in link therewith.

In this case, among other sequences other than the specific sequence used in the first symbol, a sequence having a lowest cross-correlation with the specific sequence used in the first symbol may be used in a second symbol in a sequence set/group/table.

Alternatively, according to another criterion, two sequences may be made into one pair and predetermined (or defined) to be used in two concatenated OFDM symbols or the eNB may configure (or indicate) the sequence pair to the UE.

It is conceivable that independent computer generation sequences (CGSs) are used with respect to a first (1^(st)) DMRS symbol and a second (2^(nd)) DMRS symbol, respectively.

To this end, 30 CGSs are defined per symbol to define a total of 60 CGSs. In this case, 30 sequences may be considered as one sequence set.

In this case, it may be considered that the sequence set is defined so that the cross-correlation between two sequence sets is minimized. Alternatively, when the specific sequence is used in the first (1^(st)) DRMS symbol, a specific sequence and/or specific sequence sub-set (some of 30 sequences) to be used in the second (2^(nd)) DMRS symbol may be used.

8-PSK is used with respect to length-6 CGS.

In the case of PUSCH having one OFDM symbol DMRS and pi/2 BPSK modulation, one of the following alternatives is selected.

Alternative 0: Only a single DMRS is supported (one comb is used).

Alternative 1: One DMRS port is supported per comb (in a total of 2 ports).

Alternative 2: Two DMRS ports are supported per comb (in a total of 4 ports).

Hereinafter, an additional embodiment for Method 2 described above will be described through Method 2-1.

(Method 2-1)

Method 2-1 relates to relaxing the cyclic auto-correlation performance in Method 2 above. A sequence having a better PAPR characteristic may be used through relaxation and there may be a method in which the number of sequences constituting one sequence set is significantly increased to selectively/adaptively use the sequences. In Method 2, when the condition of Method 2 is relaxed under a condition in which the PAPR is 2.3 dB or less at the time of using the FDSS filter and the PAPR is 3.2 dB or less at the time of not using the FDSS filter, sequences presented in Table 13 below may be obtained. It is proposed that one or more some sequences or all sequences of the sequences presented in Table 13 above are used as the DFT-spread-OFDM based DMRS sequence.

TABLE 13 PAPR [dB] PAPR [dB] Sequence under FDSS filter without FDSS index u Sequences ϕ_(u) (n) [0.28 1.0 0.28] filter 1. −7 −7 −5 1 −5 5 2.2775 1.7169 2. −7 −7 −3 3 −1 −7 2.2833 2.9253 3. −7 −7 3 −1 5 −7 2.2833 2.9253 4. −7 −5 −7 −3 3 −1 1.9710 2.9166 5. −7 −5 −7 −3 5 1 2.1099 2.8312 6. −7 −5 −7 −1 5 1 2.0214 3.0285 7. −7 −5 −5 −1 −7 3 2.2728 1.7325 8. −7 −5 −1 −7 3 1 2.1081 2.4678 9. −7 −5 −1 5 1 −5 1.9058 2.4184 10. −7 −5 −1 5 1 −3 2.1617 2.5641 11. −7 −5 −1 7 5 −1 2.2544 2.9451 12. −7 −5 3 −1 5 −5 1.9966 3.1184 13. −7 −5 3 −1 7 −5 2.0883 2.8683 14. −7 −5 3 1 −5 5 2.1140 2.4372 15. −7 −5 3 7 5 −1 1.9282 3.0378 16. −7 −5 5 −1 −3 1 1.9385 3.1365 17. −7 −5 5 −1 −3 5 2.2274 2.8838 18. −7 −5 5 1 7 −5 1.9941 2.8754 19. −7 −5 7 3 −1 5 2.1652 2.6502 20. −7 −3 −7 −3 1 −1 1.4462 3.1777 21. −7 −3 −7 3 1 5 1.9850 2.2841 22. −7 −3 −5 −3 5 1 2.1099 2.8312 23. −7 −3 −5 −3 7 3 1.9710 2.9166 24. −7 −3 −5 5 −1 1 1.9500 3.1462 25. −7 −3 −5 5 1 3 2.2957 3.1261 26. −7 −3 −5 5 1 5 1.9675 2.3452 27. −7 −3 −5 7 −5 −1 1.7721 2.9294 28. −7 −3 −3 −3 7 3 2.2914 3.0354 29. −7 −3 −3 −1 5 −1 2.2728 1.7325 30. −7 −3 −1 −5 7 3 2.1387 2.6592 31. −7 −3 −1 −3 7 3 1.9133 2.5031 32. −7 −3 −1 7 5 −1 2.1289 2.4781 33. −7 −3 1 −5 −1 −3 1.7721 2.9294 34. −7 −3 1 −3 −5 −1 1.7809 2.8344 35. −7 −3 1 −3 1 −5 1.4597 3.1244 36. −7 −3 1 −3 7 5 1.9675 2.3452 37. −7 −3 1 −1 −7 5 1.9850 2.2841 38. −7 −3 1 5 1 −5 1.9558 2.3581 39. −7 −3 1 5 1 −3 0.7839 3.0103 40. −7 −3 1 5 3 −3 1.9558 2.3581 41. −7 −3 3 −1 −5 7 2.1652 2.6502 42. −7 −3 5 3 −3 7 2.2274 2.8838 43. −7 −3 5 7 1 −5 1.9282 3.0378 44. −7 −3 7 −5 −1 −5 1.7592 2.9005 45. −7 −3 7 1 −1 7 2.1140 2.4372 46. −7 −3 7 1 7 −7 2.2775 1.7169 47. −7 −3 7 5 −7 −3 1.4597 3.1244 48. −7 −1 −7 3 7 7 2.2728 1.7325 49. −7 −1 −5 3 5 3 2.0214 3.0285 50. −7 −1 −5 5 3 5 1.9133 2.5031 51. −7 −1 −5 5 5 5 2.2914 3.0354 52. −7 −1 −5 5 7 5 1.9710 2.9166 53. −7 −1 −5 7 3 5 2.1387 2.6592 54. −7 −1 −3 −1 7 3 2.0214 3.0285 55. −7 −1 1 −3 7 5 2.2957 3.1261 56. −7 −1 1 1 5 −1 2.2775 1.7169 57. −7 −1 3 1 3 −3 1.9941 2.8754 58. −7 −1 3 3 3 −3 2.2833 2.9253 59. −7 −1 3 5 1 −3 2.1617 2.5641 60. −7 −1 3 5 3 −3 1.9058 2.4184 61. −7 −1 5 3 5 −3 1.9966 3.1184 62. −7 1 −3 3 −7 7 2.0214 3.0285 63. −7 1 −3 5 −7 7 2.1099 2.8312 64. −7 1 −3 5 7 5 2.1099 2.8312 65. −7 1 −1 −7 3 7 2.1081 2.4678 66. −7 1 3 −3 7 5 1.9500 3.1462 67. −7 1 3 1 5 −3 2.0805 2.9022 68. −7 1 3 1 7 −3 1.9909 3.1248 69. −7 1 3 7 1 −5 2.1289 2.4781 70. −7 1 5 3 −3 7 1.9385 3.1365 71. −7 1 5 3 5 −3 2.0883 2.8683 72. −7 1 5 7 1 −5 2.2544 2.9451 73. −7 3 −3 −5 −1 7 1.9282 3.0378 74. −7 3 −3 −5 3 5 2.1289 2.4781 75. −7 3 −3 −5 3 7 2.2544 2.9451 76. −7 3 −3 −1 3 −5 2.2274 2.8838 77. −7 3 −3 −1 7 −5 1.9385 3.1365 78. −7 3 −3 1 1 3 2.2775 1.7169 79. −7 3 −3 1 3 −5 2.1140 2.4372 80. −7 3 −3 3 5 5 2.2728 1.7325 81. −7 3 −1 3 7 −5 1.9850 2.2841 82. −7 3 −1 5 −7 −5 1.9133 2.5031 83. −7 3 −1 5 −7 7 1.9710 2.9166 84. −7 3 1 −7 −3 −1 2.2631 2.9654 85. −7 3 1 3 7 −3 1.9091 2.5047 86. −7 3 1 5 −7 5 1.4462 3.1777 87. −7 3 1 5 −5 −3 2.2957 3.1261 88. −7 3 1 5 −3 −1 1.9500 3.1462 89. −7 3 5 −7 1 −1 2.2631 2.9654 90. −7 3 5 −3 1 −1 1.9500 3.1462 91. −7 3 5 3 7 −3 1.9801 2.9603 92. −7 3 7 −5 7 5 1.7721 2.9294 93 −7 3 7 3 7 −5 1.4609 3.1778 94. −7 3 7 5 1 5 1.7809 2.8344 95. −7 5 −7 −3 −5 5 1.4597 3.1244 96. −7 5 −7 −3 1 −3 0.7788 2.9581 97. −7 5 −7 −3 7 −5 1.7809 2.8344 98. −7 5 −5 −1 −3 −1 1.9801 2.9603 99. −7 5 −5 −1 1 −3 2.1652 2.6502 100. −7 5 −5 −1 1 −1 1.9091 2.5047 101. −7 5 −5 1 −1 1 1.9909 3.1248 102. −7 5 −3 −1 −3 1 2.0883 2.8683 103. −7 5 −3 −1 −3 3 1.9966 3.1184 104. −7 5 −3 1 −1 1 2.0805 2.9022 105. −7 5 −1 −3 −1 3 1.9058 2.4184 106. −7 5 −1 −3 1 5 1.9558 2.3581 107. −7 5 −1 −1 −1 3 2.2833 2.9253 108. −7 5 −1 1 −1 3 1.9941 2.8754 109. −7 5 1 −3 −1 3 2.1617 2.5641 110. −7 5 1 −3 1 5 0.7839 3.0103 111. −7 5 1 −3 3 7 2.1617 2.5641 112. −7 5 1 3 7 −3 2.1652 2.6502 113. −7 5 1 5 −7 −3 0.7788 2.9581 114. −7 5 1 7 −5 −3 2.1387 2.6592 115. −7 5 3 7 1 5 1.7721 2.9294 116. −7 5 7 −5 1 −3 2.1387 2.6592 117. −7 5 7 −3 1 −1 2.2957 3.1261 118. −7 7 −7 −3 3 −1 1.9133 2.5031 119. −7 7 −5 −1 −5 −1 1.4609 3.1778 120. −7 7 −5 −1 3 −1 1.9850 2.2841 121. −7 7 −5 1 −3 7 1.9941 2.8754 122. −7 7 −5 3 −1 7 2.0883 2.8683 123. −7 7 −5 3 5 −1 1.9385 3.1365 124. −7 7 −5 5 −7 −3 1.7809 2.8344 125. −7 7 −3 3 −1 7 1.9966 3.1184 126. −7 7 −1 1 5 −1 2.1140 2.4372 127. −7 7 −1 3 5 −1 2.2274 2.8838 128. −7 7 1 −5 −3 1 2.2544 2.9451 129. −7 7 1 −5 −3 5 1.9282 3.0378 130. −7 7 1 −5 −1 1 2.1289 2.4781 131. −7 7 1 −3 1 5 1.9558 2.3581 132. −7 7 1 −3 3 7 1.9058 2.4184 133. −7 7 1 5 1 5 1.4597 3.1244 134. −7 7 3 7 −5 5 1.7592 2.9005

Table 13 shows one example of the proposed 8-PSK based sequence set (length-6) and the modulation symbols are generated by s_(u)(n)=e^(jϕ) ^(u) ^((n)π/8). The PAPR performance is evaluated in the DFT-s-OFDM system having the Comb-2 type DMRS for one RB.

(Method 6)

In Method 6, it is characterized in that when frequency-RE is used as Comb-2 type in the frequency domain, a time-axis signal is shown repeatedly twice. Accordingly, in order to transmit the length-6 DMRS sequence in a step before DFT in the DFT-spread-OFDM system, the length-6 sequence needs to be used by considering that the length-6 sequence is repeated twice in the time domain. Accordingly, in a (pre-)DFT procedure, the length-6 sequence should be used to be repeated twice.

$\begin{matrix} {x_{2N} = {D_{DFT}\begin{bmatrix} s_{N} \\ s_{N} \end{bmatrix}}} & \left\lbrack {{Equation}\mspace{14mu} 10} \right\rbrack \end{matrix}$

Here, D_(DFT) represents a Discrete Fourier Transform (DFT) matrix.

s_(N)=[s₁, s₂, . . . , s₆] represents a 6×1 vector (one length-6 sequence) and each element s_(i) represents the M-PSK/M-QAM symbol.

x_(2N) represents a 12×1 vector which is a frequency-domain signal after DFT processing of

$\begin{bmatrix} s_{N} \\ s_{N} \end{bmatrix}.$

In this case, when the DMRS sequence is transmitted by the DFT-spread-OFDM scheme, a DFT computation is performed by using a form in which the length-6 sequence is repeated twice in the pre-DFT step as mentioned above at the time of transmitting a single-port DMRS.

When the number of DMRS ports is 2, a specific DMRS port may be configured in a comb-2 type in which a frequency offset is 0 and the other DMRS port may be configured in a comb-2 type in which the frequency offset is 1. In this case, when the length-6 sequence to be used in the second DMRS port and the length-6 sequence to be used in the first DMRS port are mapped to a frequency axis through a (pre-)DFT process as shown in Equation 8 above, both the length-6 sequences are allocated to a Comb-2 structure (comb-2 structure in which the frequency offset is 0) having the same frequency offset. Accordingly, when both DMRS ports are used, a shifting operation on the frequency axis may be additionally required. However, in this case, since the additional shifting operation is required, the two-port DMRS sequence may be transmitted/configured to single symbols through different (even and odd) Comb-2 by the following scheme.

Sequence transmitted to first DRMS port: s_(N) ⁽¹⁾ϵC^(6×1) (6×1 vector)

Sequence transmitted to second DRMS port: s_(N) ⁽²⁾ϵC^(6×1) (6×1 vector)

The sequence may be a different sequence or the same sequence selected in the length-6 DMRS sequence table.

In order to transmit the sequences transmitted in both DMRS ports through different Comb-2, the sequence transmitted to the first DRMS port is multiplied by the DFT matrix in the form of

$\quad\begin{bmatrix} s_{N}^{(1)} \\ s_{N}^{(1)} \end{bmatrix}$

for the (pre-)DFT processing (the DFT processing is performed) and in the case of the second DMRS port, the (pre-)DFT processing may be performed in the form of

$\quad{\begin{bmatrix} s_{N}^{(2)} \\ {- s_{N}^{(2)}} \end{bmatrix}\mspace{14mu}{{and}/{or}}\mspace{20mu}{\quad{\begin{bmatrix} {- s_{N}^{(2)}} \\ {+ s_{N}^{(2)}} \end{bmatrix}.}}}$

Consequently, for the sequence transmitted to the second DMRS port, even though additional shifting processing is not performed after DFT processing, a non-zero value is configured/transmitted to six odd-numbered REs as the frequency offset is set to 1 and 0 or the non-zero value is mapped/transmitted to six even-numbered REs.

For first DMRS port:

$\begin{matrix} {x_{2N} = {D_{DFT}\begin{bmatrix} s_{N} \\ s_{N} \end{bmatrix}}} & \left\lbrack {{Equation}\mspace{14mu} 11} \right\rbrack \end{matrix}$

For second DMRS port:

$\begin{matrix} {x_{2N} = {{{D_{DFT}\begin{bmatrix} s_{N} \\ {- s_{N}} \end{bmatrix}}\mspace{14mu}{or}\mspace{20mu} x_{2N}} = {D_{DFT}\begin{bmatrix} {- s_{N}} \\ s_{N} \end{bmatrix}}}} & \left\lbrack {{Equation}\mspace{14mu} 12} \right\rbrack \end{matrix}$

In the scheme, the sequences transmitted to both ports, respectively are orthogonal to each other on the frequency axis even though time-domain OCC is not configured at the (pre-)DFT end, the sequences are distinguished. In other words, different sequences may be mapped (or configured or transmitted) to both DMRS ports.

For reference, the first and second DMRS ports mentioned in the present disclosure mean different DMRS ports and are independent of a DMRS port index.

In Methods 2, 2-1, and 2-2 mentioned above, a frequency RE mapping matrix is multiplied after DFT in order to map two antenna ports (e.g., two DMRS antenna ports) onto the frequency axis through two Comb-2 structures.

In Methods 2, 2-1, and 2-2, two different antenna ports may be considered and a case may be considered in which Method 6 above is used for Comb-2 type frequency RE mapping for each antenna port.

In other words, referring to Equations 11 and 12 of Method 6, length-6 sequences transmitted to first and second antenna ports after pre-DFT are allocated to a frequency RE in which the frequency offset is “0” and a frequency RE in which the frequency offset is “1”, respectively. By considering Method 6 above, Method 2-3 below is proposed similarly to Method 2-2.

Hereinafter, an additional embodiment for Method 2 described above will be described through Method 2-3.

(Method 2-3)

Method 2-3 relates to a method in which all sequences presented in Table 15 are used, in which each sequence element is constituted by 15-(Phase Shift Keying (PSK) symbols and the length is 6 or some of the sequences presented in Table 15 are used as uplink PUSCH and/or PUCCH DMRS sequences.

The proposed sequences may be used for Discrete Fourier Transform Spread Orthogonal Frequency Division Multiplexing (DFT-s-OFDM) and/or Cyclic Prefix OFDM (CP-OFDM). In the case of Method 2-3, since N=6 and M=8, the total number of sequences which may be considered is 8⁶. Here, a total of two antenna ports (e.g., two DMRS antenna ports) are considered and each antenna port is subject to Frequency Division Multiplexing (FDM) in the form of Comb-2.

In this case, in order to map the length-6 sequences of two antenna ports through the Comb-2 structure, like equation 11 and equation 12 of method 6 above, the length-6 sequence of the first port is repeated twice and the length-6 sequence of the second port is repeatedly output by changing a sign.

A rule of generating (or choosing or using) K (>0) some sequences among a total of sequences is as follows. Here, K=45 is assumed.

A main feature of the proposed sequence is that two antenna ports subject to FDM in the form of Comb-2 show a low PAPR characteristic of 2.2 dB or less when the FDSS filter (FDSS filter in which the time-domain response is [0.28 1 0.28]) is applied and a low PAPR characteristic of 2.9 dB even when the FDSS filter is not applied.

Additionally, the proposed sequence has the following characteristics.

-   -   The proposed sequence has a characteristic that the maximum         cyclic auto-correlation is equal to or smaller than         approximately 0.2357 in +1 and −1 cyclic auto-correlation.     -   The proposed sequence has a characteristic that the maximum         cyclic auto-correlation is equal to or smaller than         approximately 0.85 in +3, +2, +1, −1, −2, and −3 cyclic         auto-correlation lags.     -   Among K chosen sequences, all available cyclic shifted forms of         a specific length-N sequence may be regarded as the same         sequence. Accordingly, among K chosen sequences, no specific         sequence is the same as an available cyclic shifted form of         another sequence.

TABLE 14 PAPR PAPR PAPR PAPR [dB] [dB] [dB] [dB] under under without without FDSS FDSS FDSS FDSS filter filter filter filter index for the for the for the for the u Sequences ϕ_(u) (n) 1^(st) port 2^(nd) port 1^(st) port 2^(nd) port 1 −7 −7 3 7 −5 5 1.5699 2.0795 2.6332 2.2218 2 −7 −5 −1 −7 5 −1 2.1235 1.9930 2.8984 2.1349 3 −7 −5 −1 5 1 −5 1.9058 1.8246 2.4184 1.6003 4 −7 −5 −1 5 1 −3 2.1617 2.0738 2.5641 2.4144 5 −7 −5 5 1 −7 −3 1.8689 1.9449 2.1792 1.6280 6 −7 −5 5 1 −5 −3 1.7645 1.9261 2.4494 1.3633 7 −7 −5 5 1 −5 −1 1.0700 1.9930 2.4043 2.1349 8 −7 −5 5 1 7 −5 1.9941 1.4471 2.8754 2.2516 9 −7 −3 −1 −7 5 −1 2.0060 1.9261 2.5274 1.3633 10 −7 −3 −1 3 −1 −5 2.0580 2.0726 2.8206 2.4402 11 −7 −3 −1 5 1 −5 2.1968 1.8320 1.7589 1.7234 12 −7 −3 −1 7 3 −3 1.7644 2.1163 2.1336 2.1126 13 −7 −3 1 −5 7 3 1.3980 1.9740 1.4299 1.8975 14 −7 −3 1 7 3 −1 1.4188 1.9609 1.3790 1.9672 15 −7 −3 5 1 −5 −3 1.8753 1.9562 2.1527 1.6412 16 −7 −3 7 3 −3 −1 1.0858 1.9763 2.4351 2.1548 17 −7 −3 7 3 −1 3 1.5866 2.0603 2.7305 2.8314 18 −7 −1 3 −7 5 3 2.0574 1.9619 2.8329 2.1978 19 −7 3 −1 −7 −5 −1 2.0154 1.9050 2.4374 1.3575 20 −7 3 −1 −7 −3 −1 2.1603 1.9619 2.8474 2.1978 21 −7 3 −1 5 −7 −5 1.9133 1.8229 2.5031 1.6731 22 −7 3 −1 5 7 −5 2.1908 1.8229 1.7415 1.7723 23 −7 3 −1 7 −7 −3 1.7619 2.1268 2.1823 2.0435 24 −7 3 −1 7 −5 −3 1.8269 1.9568 2.8823 1.1442 25 −7 3 5 7 −5 5 1.6755 1.9763 2.4312 2.1548 26 −7 3 7 −7 −5 5 1.6696 1.9930 2.3269 2.1349 27 −7 3 7 −5 5 5 1.5402 2.0839 2.5659 2.1980 28 −7 3 7 7 −5 5 1.4371 1.6616 2.4208 1.9350 29 −7 5 −7 −3 3 −1 1.2715 2.0482 2.2025 2.8895 30 −7 5 −7 −3 7 3 2.1954 2.0738 2.8917 2.4144 31 −7 5 −5 −1 3 −1 1.3007 2.0603 2.2771 2.8314 32 −7 5 −3 3 1 5 2.0901 2.1403 2.8433 2.7642 33 −7 5 1 −5 −1 3 1.4188 1.9609 1.3790 1.9672 34 −7 5 1 5 7 −5 2.0363 2.0738 2.7382 2.4144 35 −7 5 1 7 −5 −3 2.1387 2.0726 2.6592 2.4402 36 −7 5 1 7 −5 −1 1.3980 1.9740 1.4299 1.8975 37 −7 7 −5 1 −3 7 1.9941 1.7210 2.8754 2.3917

Table 14 shows one example of the proposed 8-PSK based sequence set (length-6) and the modulation symbols are generated by s_(u)(n)=e^(jϕ) ^(u) ^((n)π/8). The PAPR performance is generated in the DFT-s-OFDM system having Comb-2 type DMRS for one resource block (RB) (TS 38.211, TS 38.214, and TS 38.331 are referred to for Comb-2 type DMRS).

-   -   The modulation symbols are generated by

${s_{u}(n)} = {e^{\frac{j\;{\phi_{u}{(n)}}\pi}{8}}.}$

-   -   u: Sequence index     -   n: Element index of each sequence)     -   The applied FDSS filter corresponds to the time-domain response         of [0.28 1.0 0.28].     -   An IFFT size is 64 and a DFT size is 12.

Some or all of the sequences presented in Table 14 above may be used.

Further, some or all of the sequences presented in Table 14 above and sequences (having different characteristics) not presented in Table 14 above table and one sequence set may be constituted and used. It may be considered that such a constitution as an extension (or application) of the method proposed in the present disclosure is included in the spirit of the method proposed in the present disclosure.

FIG. 19 illustrates PAPR performance for a proposed set of length-6 sequences in which elements of each sequence are constituted by 8-PSK symbols.

Low-PAPR Sequence Generation Type 2

Additionally, low-PAPR sequence generation type 2 will be described in brief.

A low-PAPR sequence r_(u,v) ^((α,δ))(n) may be defined by a base sequence r _(u,v)(n) according to Equation 13 below.

r _(u,v) ^((α,δ))(n)= r _(u,v)(n),0≤n≤M  [Equation 13]

Here, M=mN_(sc) ^(RB)/2^(δ) represents the length of the sequence. Multiple sequences are defined from a single base sequence through different values of α and δ.

Base sequences r _(u,v)(n) are divided into groups and here, uϵ{0, 1, . . . , 29} represents a group number and v represents a base sequence number in the group. Each group includes one base sequence v=0 having a length of M=mN_(sc) ^(RB)/2^(δ), ½≤m/2^(δ). A sequence r _(u,v)(0), . . . , r _(u,v)(M−1) is defined by Equation 14 below.

$\begin{matrix} {{{{\overset{\sim}{r}}_{u,v}(n)} = {\frac{1}{\sqrt{M}}{\sum\limits_{i = 0}^{M - 1}{{{\overset{\sim}{r}}_{u,v}(i)}e^{{- j}\frac{2{\pi n}}{M}}}}}}{{n = 0},\ldots\;,{M - 1}}} & \left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack \end{matrix}$

Here, a definition of r _(u,v)(i) depends on the sequence length.

Low-PAPR sequence generation type 2 above may be divided into (1) sequences having length 30 or larger and (2) sequences having a length less than 30.

The sequences having the length less than 30 will be described in brief.

Sequences of length less than 30

The sequence {tilde over (r)}_(u,v)(i) is given by Equation 15 below for M=6.

{tilde over (r)} _(u,v)(i)=e ^(jφ(i)π/8),0≤i≤M−1  [Equation 15]

Here, a value of φ(i) may be given as shown in the above described tables.

The sequence {tilde over (r)}_(u,v)(i) is obtained by complex value modulation symbols due to π/2-BPSK modulation for Mϵ{12, 18, 24}.

Contents regarding low-PAPR sequence generation type 2 above may be applied to the methods proposed in the present disclosure, which are described above.

The methods, embodiments, and descriptions for implementing the proposal in the present disclosure, which are described above may be separately applied or one or more methods, embodiments, and descriptions may be combined and applied.

FIG. 20 is a flowchart illustrating an example of a method for transmitting a demodulation reference signal for uplink data in the present disclosure.

More specifically, a terminal receives, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for uplink has been enabled, in S2010.

The terminal generates a low peak to average power ratio (PAPR) sequence based on a length-6 sequence, in S2020.

The terminal generates a sequence used for the demodulation reference signal based on the low PAPR sequence, in S2030.

The terminal transmits, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, in S2040.

Here, the length-6 sequence may have an 8-phase shift keying (PSK) symbol as each element of the sequence.

The length-6 sequence may be determined by e^(jφ(i)π/8), and the i may be an index of elements of the length-6 sequence.

Here, a value of the φ(i) may comprise (−1 −7 −3 −5 −1 3), (−7 3 −7 5 −7 −3), (5 −7 7 1 5 1), (−7 3 1 5 −1 3), (−7 −5 −1 −7 −5 5), (−7 1 −3 3 7 5) and (−7 1 −3 1 5 1).

A sequence which is cyclic shifted for the φ(i) may be a same sequence with the φ(i).

A number of possible value of the φ(i) may be 8⁶.

A value of an auto-correlation for the low PAPR sequence may be less than a specific value.

In addition, the terminal may apply a frequency domain spectrum shaping (FDSS) filter to the low PAPR sequence

The low PAPR sequence may be frequency division multiplexed (FDM) for 2 antenna ports as a Comb-2 form.

The low PAPR sequence which is used for each of the 2 antenna ports may be different for each other.

Detailed operations where the method illustrated in FIG. 20 is implemented in the wireless device are described.

A terminal may include a transceiver configured to transmit and receive radio signals; and a processor operatively coupled to the transceiver, in order to transmit a demodulation reference signal (DMRS) for uplink data in a wireless communication system.

The processor of the terminal may be configured to receive, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for uplink has been enabled, generate a low peak to average power ratio (PAPR) sequence based on a length-6 sequence, generate a sequence used for the demodulation reference signal based on the low PAPR sequence, and transmit, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal.

Example of Communication System to which Present Disclosure is Applied

Although not limited thereto, but various descriptions, functions, procedures, proposals, methods, and/or operation flowcharts of the present disclosure, which are disclosed in this document may be applied to various fields requiring wireless communications/connections (e.g., 5G) between devices.

Hereinafter, the communication system will be described in more detail with reference to drawings. In the following drawings/descriptions, the same reference numerals will refer to the same or corresponding hardware blocks, software blocks, or functional blocks if not differently described.

FIG. 21 illustrates a communication system applied to the present disclosure.

Referring to FIG. 21 , a communication system 1 applied to the present disclosure includes a wireless device, an eNB, and a network. Here, the wireless device may mean a device that performs communication by using a wireless access technology (e.g., 5G New RAT (NR) or Long Term Evolution (LTE)) and may be referred to as a communication/wireless/5G device. Although not limited thereto, the wireless device may include a robot 100 a. vehicles 100 b-1 and 100 b-2, an eXtended Reality (XR) device 100 c, a hand-held device 100 d, a home appliance 100 e, an Internet of Thing (IoT) device 100 f, and an AI device/server 400. For example, the vehicle may include a vehicle with a wireless communication function, an autonomous driving vehicle, a vehicle capable of performing inter-vehicle communication, and the like. Here, the vehicle may include an Unmanned Aerial Vehicle (UAV) (e.g., drone). The XR device may include an Augmented Reality (AR)/Virtual Reality (VR)/Mixed Reality (MR) device and may be implemented as a form such as a head-mounted device (HMD), a head-up display (HUD) provided in the vehicle, a television, a smart phone, a computer, a wearable device, a home appliance device, digital signage, a vehicle, a robot, etc. The hand-held device may include the smart phone, a smart pad, a wearable device (e.g., a smart watch, a smart glass), a computer (e.g., a notebook, etc.), and the like. The home appliance device may include a TV, a refrigerator, a washing machine, and the like. The IoT device may include a sensor, a smart meter, and the like. For example, the eNB and the network may be implemented even the wireless device and a specific wireless device 200 a may operate an eNB/network node for another wireless device.

The wireless devices 100 a to 100 f may be connected to a network 300 through an eNB 200. An artificial intelligence (AI) technology may be applied to the wireless devices 100 a to 100 f and the wireless devices 100 a to 100 f may be connected to an AI server 400 through the network 300. The network 300 may be configured by using a 3G network, a 4G (e.g., LTE) network, or a 5G (e.g., NR) network. The wireless devices 100 a to 100 f may communicate with each other through the eNB 200/network 300, but may directly communicate with each other without going through the eNB/network (sidelink communication). For example, the vehicles 100 b-1 and 100 b-2 may perform direct communication (e.g., Vehicle to Vehicle (V2V)/Vehicle to everything (V2X) communication). Further, the IoT device (e.g., sensor) may perform direct communication with other IoT devices (e.g., sensor) or other wireless devices 100 a to 100 f.

Wireless communications/connections 150 a, 150 b, and 150 c may be made between the wireless devices 100 a to 100 f and the eNB 200 and between the eNB 200 and the eNB 200. Here, the wireless communication/connection may be made through various wireless access technologies (e.g., 5G NR) such as uplink/downlink communication 150 a, sidelink communication 150 b (or D2D communication), and inter-eNB communication 150 c (e.g., relay, Integrated Access Backhaul (IAB). The wireless device and the eNB/the wireless device and the eNB and the eNB may transmit/receive radio signals to/from each other through wireless communications/connections 150 a, 150 b, and 150 c. For example, the wireless communications/connections 150 a, 150 b, and 150 c may transmit/receive signals through various physical channels. To this end, based on various proposals of the present disclosure, at least some of various configuration information setting processes, various signal processing processes (e.g., channel encoding/decoding, modulation/demodulation, resource mapping/demapping, etc.), a resource allocation process, and the like for transmission/reception of the radio signal may be performed.

Example of Wireless Device to which Present Disclosure is Applied

FIG. 22 illustrates a wireless device which may be applied to the present disclosure.

Referring to FIG. 22 , a first wireless device 100 and a second wireless device 200 may transmit/receive radio signals through various wireless access technologies (e.g., LTE and NR). Here, the first wireless device 100 and the second wireless device 200 may correspond to a wireless device 100 x and an eNB 200 and/or a wireless device 100 x and a wireless device 100 x of FIG. 21 .

The first wireless device 100 may include one or more processors 102 and one or more memories 104 and additionally further include one or more transceivers 106 and/or one or more antennas 108. The processor 102 may control the memory 104 and/or the transceiver 106 and may be configured to implement descriptions, functions, procedures, proposals, methods, and/or operation flows disclosed in the present disclosure. For example, the processor 102 may process information in the memory 104 and generate a first information/signal and then transmit a radio signal including the first information/signal through the transceiver 106. Further, the processor 102 may receive a radio signal including a second information/signal through the transceiver 106 and then store in the memory 104 information obtained from signal processing of the second information/signal. The memory 104 may connected to the processor 102 and store various information related to an operation of the processor 102. For example, the memory 104 may store a software code including instructions for performing some or all of processes controlled by the processor 10 or performing the descriptions, functions, procedures, proposals, methods, and/or operation flowcharts disclosed in the present disclosure. Here, the processor 102 and the memory 104 may be a part of a communication modem/circuit/chip designated to implement the wireless communication technology (e.g., LTE and NR). The transceiver 106 may be connected to the processor 102 and may transmit and/or receive the radio signals through one or more antennas 108. The transceiver 106 may include a transmitter and/or a receiver. The transceiver 106 may be mixed with a radio frequency (RF) unit. In the present disclosure, the wireless device may mean the communication modem/circuit/chip.

The second wireless device 200 may include one or more processors 202 and one or more memories 204 and additionally further include one or more transceivers 206 and/or one or more antennas 208. The processor 202 may control the memory 204 and/or the transceiver 206 and may be configured to implement descriptions, functions, procedures, proposals, methods, and/or operation flows disclosed in the present disclosure. For example, the processor 202 may process information in the memory 204 and generate a third information/signal and then transmit a radio signal including the third information/signal through the transceiver 206. Further, the processor 202 may receive a radio signal including a fourth information/signal through the transceiver 206 and then store in the memory 204 information obtained from signal processing of the fourth information/signal. The memory 204 may connected to the processor 202 and store various information related to an operation of the processor 202. For example, the memory 204 may store a software code including instructions for performing some or all of processes controlled by the processor 202 or performing the descriptions, functions, procedures, proposals, methods, and/or operation flowcharts disclosed in the present disclosure. Here, the processor 202 and the memory 204 may be a part of a communication modem/circuit/chip designated to implement the wireless communication technology (e.g., LTE and NR). The transceiver 206 may be connected to the processor 202 and may transmit and/or receive the radio signals through one or more antennas 208. The transceiver 206 may include a transmitter and/or a receiver and the transceiver 206 may be mixed with the RF unit. In the present disclosure, the wireless device may mean the communication modem/circuit/chip.

Hereinafter, hardware elements of the wireless devices 100 and 200 will be described in more detail. Although not limited thereto, one or more protocol layers may be implemented by one or more processors 102 and 202. For example, one or more processors 102 and 202 may implement one or more layers (e.g., functional layers such as PHY, MAC, RLC, PDCP, RRC, and SDAP). One or more processors 102 and 202 may generate one or more protocol data units (PDUs) and/or one or more service data units (SDUs) according to the descriptions, functions, procedures, proposals, methods, and/or operation flowcharts disclosed in the present disclosure. One or more processors 102 and 202 may generate a message, control information, data, or information according to the descriptions, functions, procedures, proposals, methods, and/or operation flowcharts disclosed in the present disclosure. One or more processors 102 and 202 may generate a signal (e.g., a baseband signal) including the PDU, the SDU, the message, the control information, the data, or the information according to the function, the procedure, the proposal, and/or the method disclosed in the present disclosure and provide the generated signal to one or more transceivers 106 and 206. One or more processors 102 and 202 may receive the signal (e.g. baseband signal) from one or more transceivers 106 and 206 and acquire the PDU, the SDU, the message, the control information, the data, or the information according to the descriptions, functions, procedures, proposals, methods, and/or operation flowcharts disclosed in the present disclosure.

One or more processors 102 and 202 may be referred to as a controller, a microcontroller, a microprocessor, or a microcomputer. One or more processors 102 and 202 may be implemented by hardware, firmware, software, or a combination thereof. As one example, one or more Application Specific Integrated Circuits (ASICs), one or more Digital Signal Processors (DSPs), one or more Digital Signal Processing Devices (DSPDs), one or more Programmable Logic Devices (PLDs), or one or more Field Programmable Gate Arrays (FPGAs) may be included in one or more processors 102 and 202. The descriptions, functions, procedures, proposals, and/or operation flowcharts disclosed in the present disclosure may be implemented by using firmware or software and the firmware or software may be implemented to include modules, procedures, functions, and the like. Firmware or software configured to perform the descriptions, functions, procedures, proposals, and/or operation flowcharts disclosed in the present disclosure may be included in one or more processors 102 and 202 or stored in one or more memories 104 and 204 and driven by one or more processors 102 and 202. The descriptions, functions, procedures, proposals, and/or operation flowcharts disclosed in the present disclosure may be implemented by using firmware or software in the form of a code, the instruction and/or a set form of the instruction.

One or more memories 104 and 204 may be connected to one or more processors 102 and 202 and may store various types of data, signals, messages, information, programs, codes, instructions, and/or instructions. One or more memories 104 and 204 may be configured by a ROM, a RAM, an EPROM, a flash memory, a hard drive, a register, a cache memory, a computer reading storage medium, and/or a combination thereof. One or more memories 104 and 204 may be positioned inside and/or outside one or more processors 102 and 202. Further, one or more memories 104 and 204 may be connected to one or more processors 102 and 202 through various technologies such as wired or wireless connection.

One or more transceivers 106 and 206 may transmit to one or more other devices user data, control information, a wireless signal/channel, etc., mentioned in the methods and/or operation flowcharts of the present disclosure. One or more transceivers 106 and 206 may receive from one or more other devices user data, control information, a wireless signal/channel, etc., mentioned in the descriptions, functions, procedures, proposals, methods, and/or operation flowcharts disclosed in the present disclosure. For example, one or more transceivers 106 and 206 may be connected to one or more processors 102 and 202 and transmit and receive the radio signals. For example, one or more processors 102 and 202 may control one or more transceivers 106 and 206 to transmit the user data, the control information, or the radio signal to one or more other devices. Further, one or more processors 102 and 202 may control one or more transceivers 106 and 206 to receive the user data, the control information, or the radio signal from one or more other devices. Further, one or more transceivers 106 and 206 may be connected to one or more antennas 108 and 208 and one or more transceivers 106 and 206 may be configured to transmit and receive the user data, control information, wireless signal/channel, etc., mentioned in the descriptions, functions, procedures, proposals, methods, and/or operation flowcharts disclosed in the present disclosure through one or more antennas 108 and 208. In the present disclosure one or more antennas may be a plurality of physical antennas or a plurality of logical antennas (e.g., antenna ports). One or more transceivers 106 and 206 may convert the received radio signal/channel from an RF band signal to a baseband signal in order to process the received user data, control information, radio signal/channel, etc., by using one or more processors 102 and 202. One or more transceivers 106 and 206 may convert the user data, control information, radio signal/channel, etc., processed by using one or more processors 102 and 202, from the baseband signal into the RF band signal. To this end, one or more transceivers 106 and 206 may include an (analog) oscillator and/or filter.

Example of Signal Processing Circuit to which Present Disclosure is Applied

FIG. 23 illustrates a signal processing circuit applied to the present disclosure.

Referring to FIG. 23 a signal processing circuit 1000 may include a scrambler 1010, a modulator 1020, a layer mapper 1030, a precoder 1040, a resource mapper 1050, and a signal generator 1060. Although not limited thereto, an operation/function of FIG. 23 may be performed by the processors 102 and 202 and/or the transceivers 106 and 206 of FIG. 22 . Hardware elements of FIG. 23 may be implemented in the processors 102 and 202 and/or the transceivers 106 and 206 of FIG. 22 . For example, blocks 1010 to 1060 may be implemented in the processors 102 and 202 of FIG. 22 . Further, blocks 1010 to 1050 may be implemented in the processors 102 and 202 of FIG. 17 and the block 1060 of FIG. 22 and the block 2760 may be implemented in the transceivers 106 and 206 of FIG. 22 .

A codeword may be transformed into a radio signal via the signal processing circuit 1000 of FIG. 23 . Here, the codeword is an encoded bit sequence of an information block. The information block may include transport blocks (e.g., a UL-SCH transport block and a DL-SCH transport block). The radio signal may be transmitted through various physical channels (e.g., PUSCH and PDSCH).

Specifically, the codeword may be transformed into a bit sequence scrambled by the scrambler 1010. A scramble sequence used for scrambling may be generated based on an initialization value and the initialization value may include ID information of a wireless device. The scrambled bit sequence may be modulated into a modulated symbol sequence by the modulator 1020. A modulation scheme may include pi/2-BPSK (pi/2-Binary Phase Shift Keying), m-PSK (m-Phase Shift Keying), m-QAM (m-Quadrature Amplitude Modulation), etc. A complex modulated symbol sequence may be mapped to one or more transport layers by the layer mapper 1030. Modulated symbols of each transport layer may be mapped to a corresponding antenna port(s) by the precoder 1040 (precoding). Output z of the precoder 1040 may be obtained by multiplying output y of the layer mapper 1030 by precoding matrix W of N*M. Here, N represents the number of antenna ports and M represents the number of transport layers. Here, the precoder 1040 may perform precoding after performing transform precoding (e.g., DFT transform) for complex modulated symbols. Further, the precoder 1040 may perform the precoding without performing the transform precoding.

The resource mapper 1050 may map the modulated symbols of each antenna port to a time-frequency resource. The time-frequency resource may include a plurality of symbols (e.g., CP-OFDMA symbol and DFT-s-OFDMA symbol) in a time domain and include a plurality of subcarriers in a frequency domain. The signal generator 1060 may generate the radio signal from the mapped modulated symbols and the generated radio signal may be transmitted to another device through each antenna. To this end, the signal generator 1060 may include an Inverse Fast Fourier Transform (IFFT) module, a Cyclic Prefix (CP) insertor, a Digital-to-Analog Converter (DAC), a frequency uplink converter, and the like.

A signal processing process for a receive signal in the wireless device may be configured in the reverse of the signal processing process (1010 to 1060) of FIG. 23 . For example, the wireless device (e.g., 100 or 200 of FIG. 22 ) may receive the radio signal from the outside through the antenna port/transceiver. The received radio signal may be transformed into a baseband signal through a signal reconstructer. To this end, the signal reconstructer may include a frequency downlink converter, an analog-to-digital converter (ADC), a CP remover, and a Fast Fourier Transform (FFT) module. Thereafter, the baseband signal may be reconstructed into the codeword through a resource de-mapper process, a postcoding process, a demodulation process, and a de-scrambling process. The codeword may be reconstructed into an original information block via decoding. Accordingly, a signal processing circuit (not illustrated) for the receive signal may include a signal reconstructer, a resource demapper, a postcoder, a demodulator, a descrambler, and a decoder.

Utilization Example of Wireless Device to which Present Disclosure is Applied

FIG. 24 illustrates another example of a wireless device applied to the present disclosure.

The wireless device may be implemented as various types according to a use example/service (see FIG. 21 ). Referring to FIG. 24 , wireless devices 100 and 200 may correspond to the wireless devices 100 and 200 of FIG. 22 and may be constituted by various elements, components, units, and/or modules. For example, the wireless devices 100 and 200 may include a communication unit 110, a control unit 120, and a memory unit 130, and an additional element 140. The communication unit may include a communication circuit 112 and a transceiver(s) 114. For example, the communication circuit 112 may include one or more processors 102 and 202 and/or one or more memories 104 and 204 of FIG. 22 . For example, the transceiver(s) 114 may include one or more transceivers 106 and 206 and/or one or more antennas 108 and 208 of FIG. 22 . The control unit 120 is electrically connected to the communication unit 110, the memory unit 130, and the additional element 140 and controls an overall operation of the wireless device. For example, the control unit 120 may an electrical/mechanical operation of the wireless device based on a program/code/instruction/information stored in the memory unit 130. Further, the control unit 120 may transmit the information stored in the memory unit 130 to the outside (e.g., other communication devices) through the communication unit 110 via a wireless/wired interface or store information received from the outside (e.g., other communication devices) through the wireless/wired interface through the communication unit 110.

The additional element 140 may be variously configured according to the type of wireless device. For example, the additional element 140 may include at least one of a power unit/battery, an input/output (I/O) unit, a driving unit, and a computing unit. Although not limited thereto, the wireless device may be implemented as a form such as the robot 100 a of FIG. 21 , the vehicles 100 b-1 and 100 b-2 of FIG. 21 , the XR device 100 c of FIG. 21 , the portable device 100 d of FIG. 21 , the home appliance 100 e of FIG. 212 , the IoT device 100 f of FIG. 21 , a digital broadcasting terminal, a hologram device, a public safety device, an MTC device, a medical device, a fintech device (or financial device), a security device, a climate/environment device, an AI server/device 400 of FIG. 21 , the eNB 200 of FIG. 21 , a network node, etc. The wireless device may be movable or may be used at a fixed place according to a use example/service.

In FIG. 24 , all of various elements, components, units, and/or modules in the wireless devices 100 and 200 may be interconnected through the wired interface or at least may be wirelessly connected through the communication unit 110. For example, the control unit 120 and the communication 110 in the wireless devices 100 and 200 may be wiredly connected and the control unit 120 and the first unit (e.g., 130 or 140) may be wirelessly connected through the communication unit 110. Further, each element, component, unit, and/or module in the wireless devices 100 and 200 may further include one or more elements. For example, the control unit 120 may be constituted by one or more processor sets. For example, the control unit 120 may be configured a set of a communication control processor, an application processor, an electronic control unit (ECU), a graphic processing processor, a memory control processor, etc. As another example, the memory 130 may be configured as a random access memory (RAM), a dynamic RAM (DRAM), a read only memory (ROM), a flash memory, a volatile memory, a non-volatile memory, and/or combinations thereof.

Example of Portable Device to which Present Disclosure is Applied

FIG. 25 illustrates a portable device applied to the present disclosure.

The portable device may include a smart phone, a smart pad, a wearable device (e.g., a smart watch, a smart glass), and a portable computer (e.g., a notebook, etc.). The portable device may be referred to as a Mobile Station (MS), a user terminal (UT), a Mobile Subscriber Station (MSS), a Subscriber Station (SS), an Advanced Mobile Station (AMS), or a Wireless terminal (WT).

Referring to FIG. 25 , a portable device 100 may include an antenna unit 108, a communication unit 110, a control unit 120, a memory unit 130, a power supply unit 140 a, an interface unit 140 b, and an input/output unit 140 c. The antenna unit 108 may be configured as a part of the communication unit 110. The blocks 110 to 130/140 a to 140 c correspond to the blocks 110 to 130/140 of FIG. 24 , respectively.

The communication unit 110 may transmit/receive a signal (e.g., data, a control signal, etc.) to/from another wireless device and eNBs. The control unit 120 may perform various operations by controlling components of the portable device 100. The control unit 120 may include an Application Processor (AP). The memory unit 130 may store data/parameters/programs/codes/instructions required for driving the portable device 100. Further, the memory unit 130 may store input/output data/information, etc. The power supply unit 140 a may supply power to the portable device 100 and include a wired/wireless charging circuit, a battery, and the like. The interface unit 140 b may support a connection between the portable device 100 and another external device. The interface unit 140 b may include various ports (e.g., an audio input/output port, a video input/output port) for the connection with the external device. The input/output unit 140 c may receive or output a video information/signal, an audio information/signal, data, and/or information input from a user. The input/output unit 140 c may include a camera, a microphone, a user input unit, a display unit 140 d, a speaker, and/or a haptic module.

As one example, in the case of data communication, the input/output unit 140 c may acquire information/signal (e.g., touch, text, voice, image, and video) input from the user and the acquired information/signal may be stored in the memory unit 130. The communication unit 110 may transform the information/signal stored in the memory into the radio signal and directly transmit the radio signal to another wireless device or transmit the radio signal to the eNB. Further, the communication unit 110 may receive the radio signal from another wireless device or eNB and then reconstruct the received radio signal into original information/signal. The reconstructed information/signal may be stored in the memory unit 130 and then output in various forms (e.g., text, voice, image, video, haptic) through the input/output unit 140 c.

In the embodiments described above, the components and the features of the present disclosure are combined in a predetermined form. Each component or feature should be considered as an option unless otherwise expressly stated. Each component or feature may be implemented not to be associated with other components or features. Further, the embodiment of the present disclosure may be configured by associating some components and/or features. The order of the operations described in the embodiments of the present disclosure may be changed. Some components or features of any embodiment may be included in another embodiment or replaced with the component and the feature corresponding to another embodiment. It is apparent that the claims that are not expressly cited in the claims are combined to form an embodiment or be included in a new claim by an amendment after the application.

The embodiments of the present disclosure may be implemented by hardware, firmware, software, or combinations thereof. In the case of implementation by hardware, according to hardware implementation, the exemplary embodiment described herein may be implemented by using one or more application specific integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs), field programmable gate arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, and the like.

In the case of implementation by firmware or software, the embodiment of the present disclosure may be implemented in the form of a module, a procedure, a function, and the like to perform the functions or operations described above. A software code may be stored in the memory and executed by the processor. The memory may be positioned inside or outside the processor and may transmit and receive data to/from the processor by already various means.

It is apparent to those skilled in the art that the present disclosure may be embodied in other specific forms without departing from essential characteristics of the present disclosure. Accordingly, the aforementioned detailed description should not be construed as restrictive in all terms and should be considered to be exemplary. The scope of the present disclosure should be determined by rational construing of the appended claims and all modifications within an equivalent scope of the present disclosure are included in the scope of the present disclosure.

INDUSTRIAL APPLICABILITY

The present disclosure is described based on an example applied to the 3GPP LTE/LTE-A/NR system, but the present disclosure may be applied to various wireless communication systems in addition to the 3GPP LTE/LTE-A/NR system. 

1. A method of transmitting, by a terminal, a demodulation reference signal (DMRS) for an uplink data in a wireless communication system, the method comprising: receiving, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generating a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generating a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmitting, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-phase shift keying (PSK) symbol as each element of the sequence.
 2. The method of claim 1, wherein the length-6 sequence is determined by e^(jφ(i)π/8), wherein the i is an index of elements of the length-6 sequence.
 3. The method of claim 2, wherein a value of the φ(i) comprises (−1 −7 −3 −5 −1 3), (−7 3 −7 5 −7 −3), (5 −7 7 1 5 1), (−7 3 1 5 −1 3), (−7 −5 −1 −7 −5 5), (−7 1 −3 3 7 5) and (−7 1 −3 1 5 1).
 4. The method of claim 3, wherein, among possible value of the φ(i), a value of the φ(i) is not identical to any of a cyclic shifted value of another value of the φ(i).
 5. The method of claim 4, wherein a number of possible value of the φ(i) is
 30. 6. The method of claim 1, wherein a value of an auto-correlation for the low PAPR sequence is less than a specific value.
 7. The method of claim 1, further comprising: applying a frequency domain spectrum shaping (FDSS) filter to the low PAPR sequence.
 8. The method of claim 1, wherein the low PAPR sequence is frequency division multiplexed (FDM) for 2 antenna ports as a Comb-2 form.
 9. The method of claim 8, wherein the low PAPR sequence which is used for each of the 2 antenna ports is different for each other.
 10. A terminal for transmitting a demodulation reference signal (DMRS) for an uplink data in a wireless communication system, the terminal comprising: a transceiver configured to transmit and receive radio signals; and a processor operatively coupled to the transceiver, wherein the processor controls to: receive, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generate a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generate a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmit, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-phase shift keying (PSK) symbol as each element of the sequence.
 11. The terminal of claim 10, wherein the length-6 sequence is determined by e^(jφ(i)π/8), wherein the i is an index of elements of the length-6 sequence.
 12. The terminal of claim 11, wherein a value of the φ(i) comprises (−1 −7 −3 −5 −1 3), (−7 3 −7 5 −7 −3), (5 −7 7 1 5 1), (−7 3 1 5 −1 3), (−7 −5 −1 −7 −5 5), (−7 1 −3 3 7 5) and (−7 1 −3 1 5 1).
 13. The terminal of claim 12, wherein a sequence which is cyclic shifted for the φ(i) is a same sequence with the φ(i).
 14. The terminal of claim 10, wherein the processor further controls to: apply a frequency domain spectrum shaping (FDSS) to the low PAPR sequence.
 15. The method of claim 10, wherein the low PAPR sequence is frequency division multiplexed (FDM) for 2 antenna ports as a Comb-2 form.
 16. An apparatus comprising one or more memories and one or more processors operatively coupled to the one or more memories, wherein the one or more processors control the apparatus to: receive, from a base station, a radio resource control (RRC) signaling including control information representing that transform precoding for an uplink is enabled; generate a low peak to average power ratio (PAPR) sequence based on a length-6 sequence; generate a sequence used for the demodulation reference signal based on the low PAPR sequence; and transmit, to the base station, the demodulation reference signal based on the sequence used for the demodulation reference signal, wherein the length-6 sequence has an 8-phase shift keying (PSK) symbol as each element of the sequence.
 17. (canceled) 